Adaptive Compass and North Stabilization Drive System

1. Field of the Invention

The invention relates to a radar system adapted for shipboard or airborne operation in which a gyro compass produces signals which are used for providing stabilization of the radar display so that the radar presentation upon the display screen does not rotate along with course changes of the ship, and for providing course related outputs such as visual readout of true bearing.

2. Description of the Prior Art

Shipboard radar systems customarily include a gyro compass which produces incremental heading information to the radar system for rotation of a true bearing scale, for providing north stabilization of the radar display, and for providing true motion correction. The most common gyro compass systems have either stepper motor or synchro-types of outputs. In the case of a stepper motor, three pole sets are commonly used with either 10' or 20' bearing change per output step. Pulse amplitudes of 35, 50 and 70 volts are normally employed. Synchro outputs, which are ordinarily three-phased, produce analog outputs representing either 1.degree. or 2.degree. of bearing per 360.degree. of shaft rotation. Somewhat less common are synchro outputs representing 10.degree. of bearing per 360.degree. of shaft rotation. The rotor is driven from a supply frequency of either 50, 60 or 400 Hz with commonly used voltages of 50, 60, 62, 115, 125 and 150 volts. The stator voltages are known to range between 20 and 90 volts with 20, 24, 57, 68, 82 and 90 volts being used in the most common commercially available units.

In past practice, the gyro compass course inputs were used to drive expensive mechanical devices, such as stepping motors, resolvers, and sine-cosine potentiometers to generate a stabilized and oriented PPI display and to rotate a true bearing scale.

Common electronic circuitry to drive the mechanical linkage from a DC stepper or 360:1 or 180:1 synchro-type gyro compass input, adaptable to a wide range of rotor and stator voltages have lately become available, avoiding the need for a new design for each different configuration. An example of one version of such a system is shown in U.S. Pat. No. 4,107,007, of J. E. Bryden, filed on Jan. 5, 1976, issued on Jan. 3, 1978, and assigned to the present assignee. However, compass systems with a synchro coupling, such, for example, that the required incremental step corresponds to a precision higher than a one sixth division of a full synchro revolution, usually requires a special mechanical solution such as remote duplication of the synchro shaft rotation to drive a version of a well-known optical shaft encoder and to acquire the desired subdivision. Such duplication adds to expense and system complexity.

Shipboard gyro compass output systems are usually susceptable to signal distortions such as high amplitude common mode transients and to uncertainties in transitions on the high level phase output gyro signal lines, resulting in display jitter, erroneous bearing read outs and, therefore, the need for frequent alignments.

Accordingly, it is an object of the present invention to provide a compass drive system adaptable for use with either a DC or AC stepper-type or synchro-type gyro compass output, including situations requiring an incremental step which corresponds to a higher precision than six divisions of a full synchro revolution and to control the orientation and stabilization of the associated PPI display.

Furthermore, it is an object of the invention to provide an adaptive compass drive to extract correct shaft rotational information from the gyro compass output devices with negligeable loading of the compass output lines and under adverse conditions of electrical disturbances.

It is another object to provide an improved compass drive system having a general capability to accept the electrical input from any incremental positioning device that sets the state, or provides the amplitude, of two or more output lines as a function of the angular or linear position of the device and to permit the highest possible speed of the input drive, commensurate with the carrier frequency used in AC systems.

It is yet another object of the invention to provide an improved compass drive system which utilizes digital directional codes in the orientation and stabilization of the associated PPI display without the need for presently used expensive mechanical devices.

These as well as other objects of the invention are achieved by a radar system in which gyro output signals of the incremental positioning devices of the synchro- or stepper-type, containing shaft position information in the relative amplitudes of the phases of the output signals at any given moment of time, are preconditioned, or normalized by phase comparator means. The gyro output signals thus provide comparator output signals which contain a normallized signal which is strobed into an output register, for AC inputs near the peak of any selected carrier half cycle, for DC outputs, whenever a change occurs, thus providing a digital code at the output of said register. The output register feeds the conditioned input signals to a microprocessor which retrieves from coded present and previous values a coded change signal representing a single or multiple step change in gyro shaft position in either direction. This output is used to update a stored angular value, representing the ship's present course. A signal containing the rotation angle is fed into a D to A converter properly phased with respect to the radar timing sequence to provide analog sweep reference voltages adapted to generate a stabilized and oriented PPI display without the usual mechanical resolvers and driving motors.

More particularly, the adaptive compass drive system of the invention accepts compass outputs from a wide variety of compass stepper or synchro output devices. The phase comparator means accepts the sinusoidal carrier type envelopes of these compass synchro output devices, preferably in the form of three individual phase signals with sinusoidal envelopes, and by comparison determines points of equal carrier amplitude between any two phases. The comparators thus divide a 360.degree. synchro shaft revolution into six divisions and provide square wave output waveforms of positive and negative half cycles for a given carrier period and shaft position. A step change in the position of the compass synchro output device causes a transition in the state of the comparator output waveforms. It should be understood that for DC stepper devices, the state of the input phase lines is a DC level with respect to "zero" level. The output of the comparators, which now compare the input signals to a DC reference, are DC levels, strobed into the output register whenever a change occurs. The carrier frequency of the synchro waveforms is removed by strobing the square waves into an output register near the peak of the selected carrier half cycle, and in this manner these gyro compass output signals are considered to have been conditioned. The register now contains the major transitions up or down of the output of the phase comparator means. This information in the output register is updated once per carrier cycle for gyro compass synchro outputs. A microcomputer retrieves from the digital coded values stored in the output register and the previous digital values, stored in the computer, a coded change signal representing an incremental change in angular position of the gyro shaft, and modifies a stored angular value. The angular data, representing the PPI rotation angle, is transferred to a digital analog conversion means which, in conjunction with analog voltages representing the current antenna shaft position data, generate real time output signals, properly phased with respect to the radar timing sequence, in the format of analog reference voltages to provide the stabilized and oriented PPI display usually with auxiliary controlled information such as operator controlled electronic bearing lines, true or relative bearing marks, north reference line and the like. Accordingly, any time the gyro shaft position passes a transition point, such as from up steps or down steps, the processor corrects the accumulated position count being used to update the digital to analog converter input signals, thereby correcting the orientation of the PPI.

Other and further objects and advantages of the invention will become apparent as the description thereof progresses, reference being had to the accompanying drawings wherein:

FIG. 1 shows a block schematic diagram of an adaptive compass and north stabilization system as part of a PPI radar system in accordance with the invention;

FIGS. 2, 3 and 4 taken together are a block schematic diagram of the adaptive compass system shown in FIG. 1;

FIG. 5A including waveforms I to III are useful in understanding the operation of the invention and FIG. 5B, waveforms IV to VI show the output of the select decoder used to select one phase between transition points of equal amplitude in the waveforms of FIG. 5A;

FIG. 6 is a synchro system wherein a 360 degree synchro shaft revolution is divided into twelve increments to represent the angular position of the synchro shaft in place of the division by six utilized in FIG. 2;

FIG. 7a and 7b are block diagrams used to describe an electronic north stabilization device which can be used in connection with the description of the digital to analog converter of FIG. 4;

FIG. 8 is a three-phase stepper gyro compass output system suitable for either polarity;

FIG. 9 is a schematic diagram of a synchro-type gyro compass;

FIG. 10 is a schematic diagram of a stepper-type gyro compass;

FIG. 10a shows the polarity of the output waveforms of the gyro compass of FIG. 10; and

FIG. 11 is a block diagram of a synchro system in which finer venier division of synchro position is obtained.

Referring briefly first to FIGS. 9 and 10, there is shown respectively a synchro-type gyro compass output device 400 and a stepper-type gyro compass output device 420. In the synchro-type gyro compass 400 shown in FIG. 9, there are provided three synchro stator coils 402 positioned electrically at 120.degree. intervals around synchro rotor coil 404. In known commercially available synchro- and stepper-type gyro compasses, iron cores are used for both stator and rotor coils. However, these are not shown in FIGS. 9 and 10 for clarity. Synchro rotor coil 404 is coupled to synchro AC reference source 406. The rotor voltage is hereinafter referred to as the carrier. Voltages of the same frequency as the carrier are induced in the stator coils. Rotation of the ship and the gyro platform causes synchro rotor coil 404 to rotate within synchro stator coils 402 producing waveforms such as those shown in FIG. 5 which represent the amplitudes of the AC voltages induced in the coils as a function of the shaft position. The synchro stator coils 402, as shown in FIG. 9, are connected to the input signal conditioner of FIG. 1.

FIG 10 shows a gyro compass 420 with a stepper-type output. The rotor of the gyro compass shown in FIG. 10 is formed by a commutator 424 the shaft of which rotates as the ship rotates in heading. The commutator has three conducting rings each covering half the circumference, but displaced over 120.degree. with respect to each other. A fourth or input ring or full circumference feeds a DC voltage to the other segments. The output of the device consists of three phase or output lines 422 connected by way of brushes 423 to each segmented ring of the commutator. The phase lines .phi.A, .phi.B and .phi.C are fed into the input signal conditioner as shown in FIG. 1.

As stated above, the output voltages from any of the synchro stator coils 402 of FIG. 9 may vary depending upon the number of turns used and the amount of coupling between the stator coils and rotor coils. The frequency of synchro AC reference source 406 is not the same among all commercially available units, and a complete cycle of the output signals may represent varying amounts of heading change depending upon the exact construction of the commercial units involved.

Referring now to FIG. 1, there is shown a block diagram of an adaptive compass and north stabilization system used as part of a PPI radar system in accordance with the teachings of the present invention. Signals from the outputs of the gyro compass are coupled to an input signal conditioner regardless of whether the gyro compass output is in the form of synchro or stepper signals. For a synchro input, a reference signal is also provided. These signals, designated .phi..sub.A .phi..sub.B and .phi..sub.C from the gyro compass provide changes in the true course of the craft to the adaptive compass and north stabilization unit.

Referring again to FIG. 1, there is shown a block diagram of an adaptive compass drive system 10 used in connection with a PPI radar and embodying the present invention. Radar antenna 12 transmits a signal by means of a transmitter, not shown, and receives radar return signals which are received by receiver 14 which, in a well-known manner, converts the return signals into video signals which are applied by sweep generator 9 and deflection amplifier 11 to the gun of a CRT 15 to intensify its screen. Antenna 12 is rotated by motor 13 by way of gear train 18. The instantaneous antenna position is mechanically coupled, in this case, in a 1:1 ratio to a conventional antenna resolver 20. This resolver on line 21 receives an input reference signal in the form of a square wave generated in a positioning sampling circuit 22. The output of resolver 20 consists of two square waves on output lines 24 and 26, the amplitudes of each being modulated by the position of the resolver as determined by the rotation of antenna 12. The amplitudes of the modulated square waves are sampled by conventional position sampling circuitry 22 which connects these inputs for a short time interval to storage capacitors, not shown, which assume the present value of the amplitude of the square waves. The ratio of analog voltages on these capacitors are a measure of and contain the angular position of antenna 12 and are referred to as the x and y antenna position reference voltages. These voltages are an input to the adaptive compass and north stabilization system 10 by way of lines 100 and 101.

Referring to the adaptive compass system 10, output signals in the case of a synchro output of FIG. 9 consists of three phases .phi..sub.A, .phi..sub.B and .phi..sub.C and the reference voltage or carrier seen in FIG. 9. The three phases .phi.A, .phi.B and .phi.C are shown in FIG. 5A which represent the amplitude of the carrier for each phase. These synchro signals represent the analog rotational position of the synchro shaft and are conditioned or transposed in the input signal conditioner 30, to be described, into a digital representation, that is, the logical "0" or "1" state of each of three output lines 15, 16 and 19 forming an input data bus 32 which is connected to microcomputer 40. The digital information A, B, C on these lines is acted upon in microcomputer 40 to sense a change in the state, ie. from 0 to 1, of any of the lines. It is noted that a change in the state of these lines is caused by a change in the position of the gyro compass, and thus a change in the state of the lines signifies an increment or decrement in the rotation of the gyro shaft, and this represents a change in the course of the craft. Microcomputer 40 has previously stored a digital value representing the former shaft position. The microcomputer now updates or modifies the stored signals to reflect the change extracted from the input signals A, B and C so that the resulting stored value represents the current course of the ship.

This extraction is done as follows: Assuming the state of the data A, B and C on lines 16, 15, 19 is stored in the microcomputer, then when a change occurs, it is compared to the former state and the microcomputer determines the significance of the change as shown immediately herebelow in Table I.

For example, starting from a state ABC (110), and the incoming signal happens to be ABC (010) representing a change in the A line from a "1" to a "0", the microcomputer detects a change has occurred in the A line which corresponds according to Table I in a change of one step in a particular direction: .DELTA.A defines 1 step Up. This can be designated as a change in shaft rotation or increment of rotation to the right. This can be seen in FIG. 5A where waveform I carrying data A is positive, waveform Ii carrying data B is positive, and waveform III carrying data C is negative at the points between 150.degree. and 210.degree.. The change .DELTA.A to 0 at 210.degree. signifies a one step change to the right. For most gyro compasses, this corresponds to a step of 1/6 of a degree. Other changes, for instance .DELTA.B at 150.degree. represents one step in the opposite direction (Down). Changes .DELTA.A.DELTA.C, representing two steps Up or .DELTA.B.DELTA.C representing two steps Down, are also extracted and implemented in the microcomputer to subtract from or add to a stored angular value and then store the changed value, for later use to be described. In effect, the microcomputer stores the current course of the ship. Thus, by subtracting or adding the extracted change in bearing information, the latest or updated value of the bearing of the ship is available at all times, permitting the retrieval of current sine and cosine of the bearing.

Referring again to FIG. 1, the microcomputer is synchronized to radar timing by a pulse repetition frequency, PRF, generated in PRF generator 28 for both the receiver 14 and transmitter, not shown. The timing pulse is also applied by way of line 29 to microcomputer 40 and acts as an interrupt or timing signal. Following interrupt, the microcomputer, by way of data bus 44, feeds out to the D to A converter 50 the sine and cosine values corresponding to the stored angular value that represents the angle of the gyro compass shaft. Thus, these digital values represent the angle over which rotation and stabilization of the PPI is to occur by way of the X and Y deflection circuit. The data on bus 42 is transmitted to the various components of the D to A conversion unit 50 by means of address bus 44 and an enable line 45. The D to A conversion unit 50 comprises D to A converters for the sine and cosine data channels. These multiplying D to A converters, to be described, are fed the digital sine and cosine of the rotation angle from the microcomputer. Also feeding the D to A converters are reference inputs for the X and Y antenna position reference X and Y voltages on lines 100 and 101, respectively. These reference voltages are applied to the X and Y D to A converters as directed by the microcomputer by way of the data and address busses. The X and Y sweep reference voltages representing the rotated PPI sweep reference are generated during PPI sweep retrace time sequentially using both D to A converters for each X or Y value of the sweep reference voltage. Thus, for each X or Y sweep reference voltage to the PPI, the microcomputer directs the appropriate X and Y antenna position reference voltages over lines 100 and 101 to the reference voltage input of each D to A converter. Each D to A converter, therefore, receives the X or Y antenna position voltages as directed by a switching unit controlled by or under command of the microcomputer by way of the above data and address busses in a manner to be described in greater detail in connection with FIG. 4. The whole process of updating the PPI reference voltages and thus moving the PPI presentation occurs once per radar period during the sweep retrace time. The latest rotation angle is retrieved for each sweep. Thus, the adaptive compass unit 10 provides, without use of intermediary mechanical devices, real-time X and Y output signals phased with respect to the radar timing sequence in the format of X and Y analog reference voltages used to generate a stabilized and oriented PPI display, usually north-stabilized.

Referring now to FIG. 2, there is shown a block diagram used when the gyro compass output consists of the three-phase synchro signals .phi.A, .phi.B and .phi.C, as described in FIG. 1. These signals represent the amplitude of the AC carrier of the synchro for a given shaft position. In particular, referring to FIG. 2 and FIG. 5A, the use of a synchro as an incremental position device is possible because of the existence of a number of easily detectable conditions such as the zero amplitude of a carrier on a phase line. This is shown in FIG. 5A as the 0, 60, 120, 180, 240 and 300 degree crossover points. This divides a 360.degree. rotation of the synchro shaft into six incremental steps. The transitions on the phase lines or waveforms at these natural division points are referred to as "major transitions". Thus, when one rotation of 360.degree. of a synchro shaft corresponds to a one degree change in the course of the ship, as is often the case, the corresponding incremental step is one-sixth degree.

The invention selects as its major transition points, not the usual above-referred to zero crossings, but rather the points of equal carrier amplitude from a comparison of any two waveforms. This is seen in FIG. 5 to be at 30, 90, 150, 210, 270 and 330 degrees. This selection eliminates transient common mode disturbances on the input phase waveforms .phi.A, .phi.B and .phi.C since there is, in effect, no fixed reference level against which a comparison need be made. This selection permits the phase components, to operate at very low signal levels while permitting an accurate amplitude comparison.

As seen in FIG. 2, the three-phase signals .phi.A, .phi.B and .phi.C are fed to an attenuator 60 of input signal conditioner 12. The voltage on each phase line is attenuated by voltage divider 61, 62, voltage divider 63, 64, and voltage divider 65, 66, which attenuate the input signals on each line inasmuch as the output at the midpoint of each of the three voltage dividers is connected, respectively, to the input of three phase comparators 71, 72 and 73 of comparison unit 70. The signal input to each comparator is attenuated preferably by a factor of approximately 20 by the output at the midpoint of each voltage divider. For example, the input resistors 61, 63 and 65 are approximately 70 thousand ohms and the output divider resistors 60, 62 and 64 are approximately 3.5 thousand ohms. It should be noted that not only is the loading of each synchro input line negligible small, but also a great or wide range of input voltages can be used without modification of input signal circuitry due to the use of attenuator 60. At the same time, each phase comparator is capable of responding to a very small signal change of approximately a few millivolts between each two input phase lines to be compared, even in the presence of relatively high common mode signal interference picked up on the three phase lines from on-board transients from heavy electrical apparatus and the like.

It should be understood that the output of comparator 71, 72 and 73 is a square wave of the frequency of the carrier. Thus, for a given position of the gyro shaft, say at 160.degree. as seen in FIG. 5, the amplitude of phase A, during the positive half-cycle of the carrier, is positive (point 80). Thus, it can be seen .phi.A is greater than .phi.C (point 83), .phi.B (point 82) is greater than .phi.A, and .phi.C is smaller than .phi.B. This is seen in waveforms I, II and III of FIG. 5A which show the output of the amplitude comparison for the positive period of the carrier. For the negative period of the carrier, the outputs are inverted with respect to I, II and III shown in FIG. 5A. These respective outputs therefore are seen as square waves at the output of the three comparators 71, 72 and 73 in FIG. 2.

The outputs of the three comparators contain the position of the synchro shaft and are represented in the relative phases of square waves 78, 79 and 81. These waveforms are strobed into a conventional storage register 90 which acts as a buffer for the microcomputer. This strobe occurs preferably near the middle of either the positive or negative period of the carrier frequency applied at the reference input terminals 52 and 53. Thus, the position information is stored as a digital code in storage register 90. The strobe is generated by means of phase shift-isolator 55 and threshold comparator 67.

The synchro reference input signal at terminals 52 and 53 is a constant amplitude carrier wave, not shown, in phase with the synchro phases and therefore with the three comparator outputs. That is, the zero crossings of the synchro reference signal and those at the output of the phase comparator occur at the same time. Referring to the phase shifter and isolator 55, the current reference signal is shifted over approximately 90 degrees whereby its zero crossings now occur close to the midpoint of the carrier waveforms 78, 79 and 80. This is achieved by the RC elements in circuit with back-to-back diodes in phase shifter 55. Optical isolator 58 isolates and rectifies the phase shifted current in the circuit in a well-known manner. The output voltage of the optical isolator 58 is waveform 87 which is the rectified half wave of the current through resistor 59 and is applied to the input of conventional threshold comparator 67 where it is compared with a well-known reference source 61 to produce a square strobe pulse 86, the positive transition of which occurs approximately half way between the ccarrier zero crossings for its positive half cycle. Thus, the positive transition of the output wave 86 of the threshold comparator on line 52 strobes the latest position information at the outputs of phase comparator 70 into register 90 by way of lines 74, 75 and 76 which is the digital code provided to the microcomputer. The angular coded information of relative shaft position is held in register 90 until updated during the following carrier period which occurs at the next positive transition of waveform 86. In the absence of angular change of the synchro shaft, the coded output remains the same. The coded waveform bits 1-3 are outputs A, B and C on lines 16, 15 and 19 which form a digital code provided to the microcomputer 40 of FIG. 1 on the data bus 32 of FIG. 1.

Referring now to FIG. 3, digital signals from storage register 90 are applied to the input of interface unit 91 of the microcomputer 40 by way of data bus 32. Microcomputer 40 is implemented using components from the Motorola 6800 family of microprocessors using the following items; an interface unit 91, which is Motorola MC6821, a microprocessor MC6802 which is an 8 bit microprocessor with a clock controlled by a crystal 93. The MC6802 also contains a random access memory, not shown, which provides the temporary storage necessary for performing calculations and storage of angular data. Also included in microcomputer 40 is an MC6840 programmable timer 94 and Read Only Memory 95 which is a model MCM2716 which stores the operational sequence for the computer. Decoder 96 permits the microprocessor 92 to address the other components in well-known computer operation.

Briefly, the signals from storage register 90 of FIG. 2 are interfaced into the microprocessor 92 when the microprocessor addresses interface unit 91 and instructs it to put coded input information on the data bus 42 for processing. As instructed by the stored data in the ROM 95, called up in sequence by the microprocessor, input information containing the current digital position code is compared by the processor to the previous digital code stored in the random access memory to determine the occurrence of a single or multiple step increment or decrement in the position of the gyro input shaft. The number of steps is added to or subtracted from the previous input shaft angle, also stored in the random access memory. Thus, the current values of svnchro shaft angle, are available in the random access memory of the microcomputer, and therefore the current sine and cosine of the shaft angle are available after retrieval from a table stored in read only memory 95. In response to the radar timing pulse, the microprocessor provides the appropriate sine and cosine values by way of the data bus 42 and address lines 44 to the input of D to A conversion unit shown in FIG. 1 and FIG. 4, properly timed with respect to the radar pulse by the programmable timer 94 which times the output data from the microprocessor with respect to the pulse repetition frequency, PRF, the basic radar timing. For example, the update of the rotated PPI reference voltages occurs in real time during the retrace of the radar PPI sweep. Timer 44 acts in response to microprocessor instructions to time the output data. Accordingly, the updated sine and cosine values of the rotation angle of the PPI become available to the D to A conversion unit of FIG. 1 at the beginning of the retrace permitting the D to A unit to combine this angular data with the present X and Y antenna position to arrive at the rotated PPI X and Y sweep reference voltage to maintain a stabilized and oriented PPI presentation under conditions of changing of ship's course.

Referring now to FIG. 4 there is shown a block diagram of the digital to analog conversion circuitry. A set of two 12-bit monolithic multiplying D to A converters, one (110) for the X reference channel and the other, (120) for the Y reference channel, receive the PPI rotation angle as 12-bit digital information, and the current antenna position angle as X and Y antenna position reference voltages, which are applied by lines 100 and 101 to the D to A converters 110 and 120 at the D to A reference voltage inputs. The analog output of each D to A converter is proportional to the product of its reference voltage and the digital value entered into the converter.

The microcomputer enters the digital PPI rotation data by way of data bus 42 and directs the data to well-known buffer registers, 111 for the X channel and 121 for the X channel. The data routing is accomplished via "Device Select" unit 130 which decodes the address inputs from the computer. Select line 45 of FIG. 4 selects the digital to analog conversion unit 50 of FIG. 1, and the three lower address bits of the address bus 44 select the individual registers within unit 50 by producing dedicated clock lines 113, 123, 133 to each register. The digital inputs, thus set and applied at the inputs of the D to A converters, are the sine or cosine of the desired PPI rotation angle. The D to A's 111 and 121 are bipolar and produce negative or positive analog voltages. The antenna position X, Y reference voltages are connected to the D to A reference voltage inputs via well-known switch unit 131, which can be a well-known analog device model AD7502 and which connects the antenna position X reference voltage to be converted to either D to A converter 110 or 120 and the Y reference voltage to be connected to either D to A 120 or D to A 110, FIGS. 7A and 7B show these connections for the sake of clarity. The switch positions are controlled by the outputs of Hardware Status register 132. The status of this register is set by the microcomputer 40 by way of data bus 42 and address inputs as decoded by 130. The microcomputer, therefore, at any time directs the antenna position reference voltages as required to obtain the proper outputs from the set of D to A's to provide the PPI sweep reference voltage. The outputs of the D to A's are connected to the input of sample and hold units 112 and 122. Under control of the output of the hardware status register 132, and therefore, as directed by the microcomputer at a given time, each sample and hold unit is set to "sample" whereby the X and Y output assumes the value applied to its input, after which the unit is reset to "hold". The D to A converters providing the analog input during the "sample" period are not free for the next task, while the sample and hold unit preserves the converter output. The microcomputer precisely times the events in the digital to analog conversion unit 50 of FIG. 1 such that PPI X, Y sweep reference voltages at the output of D to A converters 112 and 122 are updated during a radar retrace period to provide the X and Y reference value for the next sweep.

FIGS. 7A and 7B show the data inputs provided to accomplish the electronic PPI stabilization and orientation to be described. FIG. 7A shows the update of the PPI sweep X reference channel. The microcomputer transfers sequentially the sine value of the rotation angle .phi. to D to A converter 111 as described above, the cosine value of the rotation angle .phi. to D to A converter 121, directs the Y antenna reference voltage (proportional to the cosine of the antenna orientation angle A) to D to A converter 111 as its reference input voltage and the X antenna reference voltage (proportional to the sine of the antenna orientation angle A) to D to A converter 121. After a settling time of about 5 microseconds, the outputs of the D to A's stabilize to the products -cos A sin .phi. for D to A 111 and -sin A cos .phi. for D to A 121. The microcomputer now directs sample and hold unit 112 to sample in a well-known manner the sum of the D to A outputs available at summing junctio 124. The output is the inverse of the term of the outputs of the D to A's, that is, sin (A+.phi.) and then directs the unit 112 to hold this value. The microcomputer now proceeds to transfer the inverse of sin .phi. to D to A converter 111, which is now free for the next task and controls the switch to exchange the antenna position reference voltages on the units 111 and 121. The outputs of the D to A's 110 and 120 now assume the values indicated in FIG. 7B. The microcomputer now directs the sample and hold unit 122 to sample in a well-known manner the sum of the D to A outputs at summing junction 125. After approximately 5 microseconds, the output of unit 122 has assumed the inverse of this value, that is cos (A+.phi.) and the microcomputer resets unit 122 to the "hold" mode. The sets of outputs are the PPI sweep X, Y reference voltages for the rotated PPI presentation. The process of updating the sweep reference voltage is accomplished within an 80 microsecond retrace period. It should be noted that the electronic PPI orientation system, contrary to the relatively slow speed of the mechanical system, permits an almost instantaneous orientation of the PPI display to the desired stabilized direction.

It should be noted that the description of the preferred embodiment above started from a synchro-type gyro compass input which in this embodiment utilizes a division of a 360.degree. synchro shaft rotation by a factor 6. The standard steps of 1/6.degree. are achieved by a compass synchro shaft coupling ratio of 360:1. Other coupling ratios such as 180:1 or 36:1 are frequently used. The present invention in a further embodiment, to be described below, resolves these coupling ratios by adding particular decoding and selection circuitry of the input signal conditioner of FIG. 2, and by selecting in a well-known manner, another sequence of instructions in the microcomputer to process the input digital code. The remaining components of the system, ie. microcomputer and digital to analog conversion units, remain the same as seen in FIG. 2.

In addition to the feature already described above, the microcomputer 40 after detection of a transition in the digital input code obtained from register 90, verifies the validity of the new input. For example, an all "zero" or all "one" input constitutes an error condition and is ignored by the microcomputer as an invalid input. Before a new position is accepted by the microcomputer as the current position, a change of the digital code in response to a change of ships course, in register 90 is confirmed for a preset number of periods before a new position is accepted by the microcomputer as the current position. This permits the removal of input jitter by adding an effective hysteresis or delay to the position data inputs received from the gyro compass.

Multiphase systems as opposed to two-phase systems, such as an incremental optical encoder, contain more information than just a single one step up or down. The synchro position input, described above, provides a three-element digital code A, B, C at the output of register 90 and is interpreted as one or two steps up or down, as previously described with respect to Table I.

This two step interpretation permits higher input data speeds, permitting the system to accept faster changes in course from the gyro compass than when a single step change is processed as in the prior art. The maximum input rotation speed that is now capable of being accepted corresponds to a 120.degree. synchro shaft position change per carrier period. For a syncro used with a 60 Hz carrier, this speed of the synchro shift can amount to a speed of 20 revolutions per second or 1200 RPM. Using a 360:1 compass synchro coupling, this amounts to a course change of 20.degree./per second. Airborne applications where faster course changes are frequently encountered use a 400 Hz carrier. The speeds are proportionally higher, such as speeds up to approximately 130.degree. course change per second.

Referring now to FIG. 6 there is shown a block diagram of the input signal conditioner wherein a 360.degree. synchro shaft revolution is divided by 12, as is the case when the gyro compass synchro shaft coupling is 180:1. Added circuitry consists of a select decoder 68 which operates on the outputs 78, 79, 81 from the phase comparators 70 by taking the "exclusive OR" between any two outputs, of waveforms I, II, or III of FIG. 5A. FIG. 5B waveforms IV, V and VI show the result, in which, for example at 60.degree., waveform I and III are both "high", to yield a zero output on waveform VI, selecting .phi.C.

The output lines 34, 35, and 36 of select decoder 68, respectively, carry waveform IV, V, and VI. Only one of these waveforms is negative due to exclusive OR decoding of unit 68 and shown in FIG. 5B. The negative portion of the waveform is used to select the particular attenuated .phi.A, .phi.B, or .phi.C signal at the input of switch unit 69 to provide on line 84b this selected and attenuated signal. Such unit can be a well-known solid state switch such as AD 7511 of Analog Devices. Each line IV, V, or VI when negative, therefore selects the input phase signal .phi.A, .phi.B, .phi.C with decreasing amplitude between major transition points. The selected attenuated phase .phi.A, .phi.B or .phi.C on line 84b is compared to a voltage level to achieve a subdivision of the synchro shaft revolution between major transition points 83B, 83D, 83E, 83F, 83G, and 83H shown in FIG. 5A. These are points of equal amplitude as previously described with respect to the operation of phase comparator of FIG. 2. Subdivision transitions are referred to as "minor transitions" below. The selection of the phase with decreasing amplitude between any two points of equal phase amplitudes, as representative of the input signal, effectively replaces the set of three individal phases with sinusoidal envelopes with a single signal with a triangular envelope formed by points 83A, 83B, and 83C shown in FIG. 5A. The selected phase is susceptable to any common mode transient disturbance on the input signal. The invention removes the common mode signal by observing that the sum of the instantaneous amplitudes of the synchro phases are always zero, providing the means to establish a virtual zero for the synchro input signal at the common point of the voltage dividers of attenuator 60. Connected by way of line 140 to level adjust unit 141, the voltage difference between this virtual zero and local ground on line 140 is proportional to the common mode transient and is used to eliminate any common mode component from the selected phase .phi.A, .phi.B or .phi.C in the level adjust unit 41. Level adjust unit 141 of FIG. 6 acts upon the selected phase signal on line 84b to adjust its level according to the value of the signal on line 140, representing the common mode voltage. The average voltage of the output signal on line 145 is now at the local ground level and thus the common mode signal is removed. This ground level referenced signal permits voltage comparison to be made in comparator 146, with respect to the local ground, thus defining the midpoint between the major transitions. The output of the comparator 146 is a square wave, the "1" OR "0" output depending on whether the amplitude of the carrier is positive or negative. For instance, in FIG. 5B, waveform VI, during its negative state selects .phi.C between points 83B and 83D. The output of comparator 146 is "1" between 83C and 83B, "0" between 83C and 83D during the positive phase of the carrier and inverted during the negative phase. This output is fed to the data register 90 and clocked into the same way as described above for the 360:1 synchro, removing the carrier. The digital code at the output of the register 90 now consists of 4 bits namely A, B, C and D to be examined by the microprocessor and processed with the transitions in the input code A, B, C, and D, to provide update of the stored position of the gyro compass and thus of the course of the ship.

Table II above shows the decision format for the microprocessor which decides whether input transitions signify one, two, three, or four steps up or down. A compass synchro coupling ratio 180:1 producing 1/6.degree. steps thus permits a compass course variation of up to 40.degree./second with a 60 Hz carrier.

FIG. 11 is a block diagram of the input signal conditioner for the case where a subdivision between major transition point greater than 2 is provided, such as for a compass synchro shaft coupling of 36:1, using nine minor transition points between adjacent major transition points thus giving a subdivision of ten minor steps between the major transitions. Slower shaft speeds can now be used to give the same 1/6.degree. step changes in course. The implementation is essentially the same as for the method described where comparator 146 adds 1 minor transition point. The comparison is local ground 151 in FIG. 6 is now replaced by a comparison with an analog voltage obtained from a four-bit D to A converter by way of line 149. The digital input to the D to A convertor 147 is provided by the microcomputer 40 by way of a buffer register register 148.

More particularly, comparator 146 determines whether the carrier amplitude of the selected phase on line 152 is within the step assigned by the microcomputer 40 by means of storing a particular step value in buffer register 148 which is then converted by D to A converter 147 into the analog comparison voltage. When the carrier amplitude on line 152 is higher than a predetermined range of voltages adjacent to the analog comparison voltage on 149, a digital output "1" occurs on increment line 155, when lower than this range of analog voltage a "1" occurs on decrement line 157. In the case where the carrier amplitude on line 152 is within this range of analog voltages, both increment and decrement lines are "0". Increment and decrement lines are strobed simultaneously with outputs A, B, and C into register 90, in the previously described manner. The digital code at the output of the register 90, now consists of 5 bits namely A, B, C, D and E, with D and E commands to the microprocessor to respectively increment or decrement the digital value strobed into buffer register 148 of FIG. 11. Thus, the outputs of register A, B, C, D and E on data bus 32 are processed in the microcomputer which raises or lowers its output to the D to A converter 147 via buffer register 148 until equalization of the two input signals to comparator 146 occurs. This determines the particular minor step which corresponds to the position of the gyro synchro shaft. When detecting a transition from one step to an adjacent step, the microcomputer adds or subtracts the step value to the present stored value representing the stored angular course of the ship. The 4-bit D to A converter permits implementation of subdivisions up to 16. The linearity of carrier envelope on line 152 between major transition points is such that a subdivision of up to 20 monotonous increments is possible. In case of misalignment, the microcomputer provides automatic position correction at the major transition points.

Referring now to FIG. 8, there is shown a block diagram of the input signal conditioner 30 for use when the gyro compass has a stepper output instead of a synchro output. For a DC stepper, the outputs of the gyro compass are .phi.A, .phi.B and .phi.C waveforms which appear on lines 422A, 422B and 422C, respectively, as shown in FIG. 10. FIG. 10A shows when each of these waveforms is positive (solid) or negative (blank). The input conditioner 30 accepts DC signals on lines 61, 63 and 65 of either polarity over a wide range of voltage levels. The attenuator and isolator 60B includes a diode bridge 170 connected across line 61 and 171 in series with optical isolator 172 which operates in a manner similar to the optical isolator 58 of FIG. 2. .phi.B and .phi.C are coupled in the same manner as .phi.A. The use of the diode bridge 170, in a well-known manner, permits signals of either polarity to be accepted with the current through the optical isolator independent of the polarity. Thus, the polarity of the source 175 of FIG. 10 can be of either polarity. Comparators 70B are now configured to compare each phase input against a preset level set by potentiometer 177. The outputs A, B, C on lines 74, 75, 76 are clocked into register 90 whenever a transition occurs as detected by transition decoder 181. For AC stepper inputs, when DC source 175 of FIG. 10 is replced by an AC source, the signal has a carrier, and the waveforms of FIG. 10A are the envelopes of the carrier amplitudes. Data is now clocked into the register 90 in the same way as described for synchros as in FIG. 2, instead of using transition decoder 181.

This completes the description of the embodiments of the invention illustrated herein. However, many modifications thereof will be apparent to persons skilled in the art without departing from the spirit and scope of the invention. Accordingly, it is intended that the invention be not limited to the particular details of the embodiments described herein except as defined by the appended claims.