Phasor Impedance Measuring Test Set

The present invention relates to test set apparatus for analyzing circuit characteristics. More particularly, it relates to phasor impedance measuring test set apparatus for use in making measurements on electrical circuits including cable transmission systems, particularly telephone communication cables.

Telephone cable and communication network circuit analysis, including performance evaluation, troubleshooting of existing installations and acceptance testing of new installations for faults, is greatly facilitated by having knowledge of the electrical impedance and return loss looking into the two-wire ports of the cable or network circuit. For example, testing of subscriber loop performance is facilitated by having knowledge of the impedance on subscriber loops looking out from the central office.

Traditionally, the telephone industry has utilized return loss measurements as the accepted characteristic to quantify the impact of impedance mismatches on the design and operation of communications circuits and networks. Return loss is a measure of the signal power that will be reflected at a point of impedance discontinuity along a transmission path. In effect, return loss in one measure of the efficiency of signal propagation at impedance mismatches. For example, if the return loss at a selected point along a transmission line is 10 dB, then one-tenth of the incident signal power that appears at that point will be reflected toward the source; that is, "returned" toward the originating point. Nine-tenths or 90% of the incident signal power will propagate beyond the point of impedance mismatch.

Return loss measurements, although related to impedance, do not give an indication of the actual impedance values occurring at the point under consideration. The telephone industry has adopted measurement of return loss in their standard methods primarily because of the lack of practical instrumentation to measure impedance directly. Existing apparatus for measuring impedance have been in the form of balanced bridges, such as the familiar Wheatstone bridge, or other similar apparatus, such as level tracer instruments. These types of apparatus have been too time-consuming to use and lacked the ability to measure impedance in the presence of unwanted signals that normally occur on practical telecommunications circuits.

Accordingly, there has developed a need for improved test set apparatus for measuring impedance, which is suitable for use in telephone cable and network circuit analysis testing.

The present invention provides a test set instrument for analyzing the characteristics of a circuit. Specifically, a test set is provided for measuring the phasor impedance of a circuit under test, and for calculating return loss of a circuit under test. As used herein, "circuit" and "circuit under test" specifically include a circuit which is balanced with respect to ground such as a loaded or unloaded cable transmission system, a telephone circuit, a telephone cable, a subscriber loop line, and telephone termination equipment.

In accordance with one aspect of the invention, the resistive and reactive components of the phasor impedance of a circuit under test are measured at a measurement frequency selected by direct entry through keyboard input. A microcomputer calculates the magnitude and phase of the measured phasor impedance, and a display readout of the phasor impedance in absolute value and phase angle is made. Alternatively, the measured resistive and reactive components of the phasor impedance are displayed.

In accordance with another aspect of the invention, a test set is provided for calculating the return loss of a circuit under test. The test set measures at a selected measurement frequency the impedance of the circuit under test. A microcomputer calculates the return loss for display. Further in accordance with this aspect of the invention, a plurality of keyboard selectable return loss calculation program subroutines, each comprising an instruction set serving upon command to instruct predetermined computation processing by the microcomputer's CPU, are stored in a memory accessed by the CPU. The stored program subroutines provide for calculation of return loss against the impedance at the measurement frequency of a predetermined RC network configuration with either predetermined or user-selected values of resistance and capacitance.

As yet another aspect of the present invention, in a test set providing for measurement of the impedance of a circuit under test, which includes a signal generator producing a test signal of a selected measurement frequency, provision is made for direct entry of a selected measurement frequency value into the test set, store and recall of a plurality of measurement frequency values, and user-controlled step changes in the test signal frequency. Entry of test parameter information, including measurement frequency values, into the test set is by keyboard input, with a memory storing entered test signal frequency information and a microcomputer controlling the signal generator.

As a further aspect of the present invention, there is provided in a test set apparatus for measuring the impedance of a balanced circuit, which includes a current source driven by a signal generator to produce measurement current flow through a circuit under test and a measurement circuit to detect the resulting voltage drop, a battery power source for supplying power individually to the current source so as to isolate it from ground and permit test set operation in the presence of externally-induced longitudinal voltages.

With the present invention, direct measurement of the phasor impedance of a telecommunications circuit can be made rapidly and in the presence of the unwanted signals that appear in practice. Further, the invention allows for the measurement of the phasor impedance of active circuits, that is, on dc-powered circuits. The invention further allows for definition of the performance of a circuit under test in terms of the return loss, while also quantifying the actual phasor impedance in terms of its resistive and reactive components. The component quantities help the user to determine the cause of good or poor return losses and, therefore, provide the user with more complete information with which to take corrective action. For example, with a test set in accordance with the present invention, it can be determined whether the impedance of a circuit under test has a high or low resistive component and to what degree the impedance reactance is capacitive or inductive. With this information, impedance compensation or "buildout" can proceed more directly than is possible with the more restrictive return loss information typically available. Also, impedance measurements can be made on circuits exhibiting high longitudinal balance. Longitudinal balance is important on telecommunications circuits because it determines the degree to which interfering signals, particularly those derived from the influence of adjacent power distribution systems, will be converted to unwanted noise or cause harm to electronic equipment attached to the telephone cables. The higher the longitudinal balance, the better. Telephone line circuits exhibiting high longitudinal balance have a high degree of symmetry in the impedance from the tip conductor to ground and from the ring conductor to ground. The present invention provides test set apparatus which is highly balanced and, therefore, will not interfere with or degrade the longitudinal balance achieved on transmission cable circuits.

The present invention, although particularly well-suited for telecommunications applications, also provides similar advantages to the electronics industry in general. This is because the present invention provides the ability to measure phasor impedances directly at any selected frequency in the audio band, even in the presence of unwanted signals, and on active circuits.

These and other aspects and capabilities of the present invention will become better understood from the detailed description of an illustrative embodiment which follows.

A written description setting forth the best mode presently known for carrying out the present invention, and of the manner of implementing and using it, is provided by the following detailed description of a preferred embodiment which is illustrated in the attached drawings wherein:

FIG. 1 is a pictorial perspective view of the test set apparatus in accordance with the present invention;

FIG. 2 is a generalized block diagram of the test set apparatus;

FIG. 3 is a simplified schematic diagram of the microcomputer portion of the test set apparatus diagrammed in FIG. 2;

FIG. 4 is a simplified schematic diagram of the signal generator of the impedance measuring portion of the test set apparatus diagrammed in FIG. 2;

FIG. 5 is a simplified schematic diagram of the measurement circuit portion of the test set apparatus diagrammed in FIG. 2;

FIG. 6 is a simplified schematic diagram of the analog-to-digital converter portion of the test set apparatus diagrammed in FIG. 2; and

FIG. 7 is a simplified schematic diagram of the keyboard and display portions of the test set apparatus diagrammed in FIG. 2.

A. Test Set Apparatus

The phasor impedance measuring test set 10 as a packaged instrument is illustrated in FIG. 1. The test set 10 includes a housing 12 having a front panel 14. On the front panel are connectors 16, 18 and 20 which serve as measurement terminals for connection to a circuit under test. In the case of a circuit under test being a telephone cable pair, connectors 16 and 18 are connectable to the tip and ring lines. Accordingly, on the front panel, connector 16 is marked T and connector 18 is marked R. Connector 20 serves as a ground connection and is marked G on the front panel.

The front panel of test set 10 also includes three displays 22, 24 and 26. Display 22 shows the frequency in Hertz during measurement. Displays 24 and 26 show either resistance and reactance in kilohms, respectively, or impedance in kilohms and phase angle in degrees, respectively.

A numeric and function selection keyboard 28 is provided on front panel 14. Keys 0-9 enter digits 0-9 into the frequency display 22. Keying the first digit stops all measurement operation and places the test set in the key entry mode. The numeric keys are also used for designating memory storage locations for entered measurement frequency values. The ENT key places the test set in the measure mode after a frequency has been keyed into the frequency display. The SEL key selects the impedance range for measurement. Successive keying of the SEL key cycles the impedance range through Low, High and Auto. Light emitting diode indicators 30, 32 and 34 are provided to indicate the range. The DM key establishes the display mode by changing the impedance display between rectangular and polar notation. Light emitting diodes 36 and 38 indicate the display mode. The Step keys 40, 42 are effective to change the test signal measurement frequency in increments or decrements of a predetermined number of Hertz. The STO key, followed by depression of a numeric key, stores the displayed measurement frequency value in a storage register. As many as ten frequencies can be stored in locations designated 0-9. The RCL key, when followed by depression of a numeric key, recalls a measurement frequency value from the designated storage register location. Additional light emitting diode indicators 44, 46 and 48 are provided on front panel 14 to show measurement mode operation, measurement frequency entry mode, and measurement frequency recall mode, respectively.

Turning now to FIG. 2, there is provided a functional block diagram of test set 10. Generally, the apparatus comprises a microcomputer 50, impedance measuring means 60, keyboard input means 70 and display 80. Microcomputer 50 includes interconnected central processing unit (CPU) 52 and memory, including read only memory (ROM) 54 and random access memory (RAM) 56. A program including a plurality of keyboard-selectable subroutines is stored in memory. The main program routine and each of the subroutine programs comprises an instruction set adapted upon command to initiate predetermined control and processing actions by the CPU. Specifically, the CPU generates digital control signals to the impedance measuring means 60 of the test set and generates digital measurement data for readout on display 80. Microcomputer 50 communicates with other portions of the test set apparatus via a system bus 58.

Impedance measuring means 60 provides for measurement, at a selected measurement frequency of the impedance of a circuit under test connected to measurement terminals 16, 18. Impedance measuring means 60 includes a signal generator 62 for generating a test signal of a selected measurement frequency. The test signal frequency is selected by entry of a measurement frequency value on the front panel keyboard 70. The generated test signal is applied to a current source 64, which produces measurement current flow through a circuit under test. A measurement circuit 66 connected to the measurement terminals detects in-phase and quadrature components of the voltage drop that results from measurement current flow, and produces voltages proportional to the resistive and reactive components of the impedance. Analog-to-digital converter apparatus converts the detected voltage drop components into digital values proportions to the resistive and reactive components of the phasor impedance of a circuit under test. The digital values are returned over bus 58 to microcomputer 50. Based upon the resistive and reactive impedance component values, microcomputer 50 calculates an absolute magnitude value and phase angle of the phasor impedance.

Keyboard 70 and display 80 have already been generally discussed in connection with the description of the front panel of test set 10. Further description of the keyboard and display sections will be provided in connection with the discussion of FIG. 7.

Turning now to FIG. 3, there is presented a more detailed diagram of microcomputer 50. An oscillator 102 provides the necessary clock input to the CPU. Another input to the CPU is the interrupt request input designated NMI. This signal input causes the CPU to go to a designated location in its operating program. Memory 54 is shown to be in the form of programmable read-only memory. ROM 54 is addressed by a 12-bit address on lines A0-A11 of 16-bit address bus 104 and by one of two enable signals generated by decoder 106. Random access memory 56 includes memory chips RAM-1 and RAM-2. Again, eight bits of address, A0-A7, from address bus 104 and an enable output from decoder 106 provide the memory chip and location selection addressing for RAMs 56A and 56B. The read and write signals (RD and WR) for RAM 56 are produced from signals generated by the CPU and combined by logic comprising OR gates 108, 110. Digital data read out of memory, writtin into memory input/output to the CPU, and input/output to other portions of the test set apparatus is carried over bus 58, which consists of eight bits, D0-D7.

Referring next to FIG. 4, the signal generator of the impedance measuring means is schematically diagrammed. A digital input control word from microcomputer 50, representing the desired test signal measurement frequency, is loaded into latches 120, 122. The desired measurement frequency is represented by a 16-bit binary number. Because bus 58 is only 8 bits, the binary measurement frequency value must be loaded into the latches in two parts. The first part is the higher order bits, which are loaded into latch 120, and the second part is the lower order bits, which are loaded into latch 122. To appropriately synchronize the loading of the two parts of the binary measurement frequency value, decoder 124 and logic comprising NOR gates 126, 128 is included. Address bits A0, A10, and A12-A15 are applied to decoder 124 which generates a binary output on output decode lines 130, 132. After data has been set up at the latch inputs and address decoder 124 has enabled the appropriate one of gates 126 or 128 for issuance of a load pulse, the WR signal is issued from CPU 52. The 16-bit binary measurement frequency value is made available from latches 120, 122 as binary Word B.

Word B is applied as one input to an accumulator circuit comprising a series of interconnected full adders 134, 136, 138, 140 and 142. Word B is added to Word A, which is a binary value stored in latches 144, 146 and 148, and made available over lines A0-A19. Word A is the previous sum output from the full adders. Latches 144, 146 and 148 are clocked by the CLK signal from oscillator 102 in FIG. 3 to load and maintain the sum output of the adders.

The accumulator capacity is 20 bits; and the frequency of CLK is 1.0485 MHz or 2.sup.20 Hz. Accordingly, if the measurement frequency value input were 1 Hz, such that the accumulator latches content increases in value by one on each clock pulse, 2.sup.20 clock pulses would be required for the accumulator content to go from zero to full count. In time, this would take one second. Similarly, if the measurement frequency input value is for 100 Hz, the accumulator content would increase from zero to full count 100 times a second.

The accumulator output, i.e., the outputs from latches 144-148, may be used to determine the instantaneous value of a sine wave at each clock pulse. While a full 20-bit accumulator is required to obtain 1 Hz resolution in the frequency input, 11 bits provides sufficient accuracy in producing the instantaneous levels of a sine wave. Accordingly, the nine least significant bits from the accumulator are not used. Also, the symmetry of a sine wave permits the use of a half-wave sine look-up table in the form of ROM 150, which has a 10-bit input and an 8-bit output.

Also available from latch 148 are quadrant and sign bits. The sign bit (SIGN) is the most significant bit from the accumulator and is in-phase with two sweeps through the addresses of the look-up table in ROM 150. That is, the sign bit is a square wave signal that corresponds in frequency and phase to the desired output signal, with each half-cycle of the output being produced by one sweep through the addresses of ROM 150. The quandrant bit (QUAD), however, is a square wave at twice the output frequency that goes through a complete cycle during each sweep through the ROM 150 addresses. The SIGN and QUAD signals are produced by feeding latch outputs A19 and A18, respectively, back to two available latch inputs, thereby introducing a time delay of one clock interval corresponding to the delay in the input latches in D/A converter 152.

The digital outputs from ROM 150 are applied as the input to an 8-bit digital-to-analog (D/A) converter 152. Level adjustment for the output of D/A converter 152 is provided by a series combination of resistor 154 and potentiometer 156. The level-adjusted output of D/A converter 152, a full-wave rectified sine wave, is applied through resistor 158 to the inverting input of operational amplifier 160. The output from D/A converter 152 is also applied to an RC network of resistor 166 and capacitor 168, which is connected to the non-inverting input of operational amplifier 160. The non-inverting input of operational amplifier 160 is also connected to a transmission gate 170 controlled by the sign bit (SIGN) from latch 148.

By control of transmission gate 170 by the sign bit, operational amplifier 160 changes in gain between an inverting gain of one and a non-inverting gain of one. Thus, the half sine wave signal (at twice the output frequency), from D/A converter 152 is inverted during one half-cycle of the output signal and not inverted in the other half-cycle. Accordingly, a sine wave output at the desired measurement frequency is produced by the signal generator in FIG. 4.

The test signal generated by the signal generator of FIG. 4 is applied to current source circuitry in FIG. 5. Specifically, the test signal is applied to an input network of resistors 172, 174 and 176. Connected to the input resistor network are transmission gates 178, 180, which are controlled by the Q output of S/R flip-flop 182. Depending upon the state of flip-flop 182, one of the transmission gates is "closed" and the other is "opened". Closure of one of the transmission gates rather than the other will alter the effective voltage divider ratio of the resistor input network, and change the level of the signal applied to buffer amplifier 184. In this manner, as will become more fully appreciated, the impedance measurement range may be changed between the high and low ranges.

The buffered signal is coupled through transformer 186 to a battery-powered current source circuit. The battery power source comprises interconnected battery banks 188, 190. Relay contacts 192, 194 controlled by relay coil 196 alternately connect the battery source between a charging circuit and the constant current source circuitry. Relay coil 196 is driven by field effect transistor 198 which receives a control input designated FLOAT.

A resistor 200 is connected across the secondary winding of transformer 186. One end of resistor 200 connects to a non-inverting buffer amplifier 202, and the other end of resistor 200 connects to relay contact 204. The output of buffer amplifier 202 is coupled by capacitor 206 to a current source built around operational amplifier 208. Op-amps 202 and 208 are part of a dual amplifier package; accordingly, both are powered from the battery power source. The signal coupled through capacitor 206 is applied via resistor 210 to the inverting input of amplifier 208 to which a feedback resistor 212 connects from the amplifier output. The output current signal is applied through output resistor 214 and protective resistor 216 to relay contact 218. Positive feedback from the output end of resistor 214 is applied to the non-inverting input of amplifier 208. The combination of negative and positive feedback is arranged to produce a constant output current, independent of the impedance through which the current may flow so long as the maximum output voltage capability of amplifier 208 is not exceeded. The closure of relay contacts 204, 218 is controlled by relay coil 220, which is in turn controlled by a field effect transistor driver 222. A signal designated OUTPUT ENABLE is applied to the gate of field effect transistor driver 222 to effect the opening and closing of the relay contacts.

As will be noted, relay contacts 204, 218 are coupled to the measurement terminals 16, 18 (i.e., the tip and ring lines) by DC blocking capacitors 224, 226. When relay contacts 204, 218 are open, capacitors 224, 226 are connected in series, respectively, with resistors 228, 230 to ground.

Connected between the battery power source and node 232 are diodes 234 and 236. These diodes provide protection against electrical transients appearing on measurement terminal 16, by routing the same into the battery.

A test signal, a sinusoidal current of the form I sin .omega.t (where .omega.=2.pi.f, and f is the measurement frequency), is thus applied to a circuit under test connected to measurement terminals 16 and 18. This establishes measurement current flow through the unknown impedance of a circuit under test. The voltage appearing across the measurement terminals is detected by a measurement circuit, with the voltage being coupled through DC blocking capacitors 240, 242, and through isolating resistors 244, 246, to a differential input/single-ended output amplifier circuit comprising operational amplifiers 248, 250. Feedback resistor 252 and a balancing potentiometer 256 are provided for amplifier circuit 248. The output from amplifier 248 is applied via resistor 258 to the inverting input of amplifier 250, which also has a feedback resistor 254 connected thereto.

The output voltage signal from the single-ended output amplifier circuit is coupled through capacitors 260, 262 to gain control amplifier circuit 264 and a unity-gain inverter circuit 266. The voltage signal coupled through capacitors 260, 262 is applied to the inverting input of an operational amplifier via input resistor 268. A feedback resistor 270 and gain adjustment potentiometer 272 establish the gain for circuit 264, which provides additional amplification to the voltage signal. The amplified output voltage signal from circuit 264 is applied by resistor 274 to the inverting input of the operational amplifier of circuit 266. A feedback resistor 276 connects between the output and the inverting input of the operational amplifier. Resistors 274 and 276 are chosen to be of equal value, thereby establishing a gain of one through circuit 266. The output of circuit 266 is, however, inverted from the output of circuit 264.

Accordingly, the output of circuit 264 may be designated as +V and the output of circuit 266 designated as -V.

The output voltage signals from circuits 264, 266 are applied to transmission gates 278, 280 and 282, 284. Specifically, the output of circuit 264 is applied to transmission gates 278, 282, and the output signal from circuit 266 is applied to transmission gates 280, 284. Transmission gates 278, 280 are controlled by the SIGN bit signal from the signal generator. As will be recalled, SIGN is a square wave signal in-phase with the signal generator sinusoidal signal output. It is to be recognized, however, that transmission gates 278, 280 conduct on opposite half cycles of the SIGN square wave. Thus, when SIGN is "high", gate 280 conducts; and when SIGN is "low", transmission gate 278 conducts.

Transmission gates 282, 284 are controlled by a signal generated by the Q output of D-type flip-flop 286. The clock input to flip-flop 286 is the quadrant signal, QUAD, from the signal generator. The D input of flip-flop 286 is SIGN. The signal from the Q output of flip-flop 286 will correspond to the SIGN square wave; however, it will be shifted by 90.degree., providing a quadrature signal with respect to the signal generator output sign and the measurement current.

The outputs of transmission gates 278, 280 are connected together, and the outputs of transmission gates 282, 284 are connected together. Accordingly, both pairs of transmission gates are connected as single-pole, double-throw electronic switches.

As will be recalled, the output from the current source to the measurement terminals may be expressed as I sin .omega.t. When this current passes through the phasor impedance, Z, of a circuit under test, the resulting voltage drop, which appears across the measurement terminals, may be expressed as:

V=IR sin .omega.t+IX cos .omega.t

where R is the resistive (or Real) component of Z, X is the reactive (or Imaginary) component, and .omega. is 2.pi.f, where f is the measurement frequency. A voltage proportional to V appears at the output of circuit 264, and a voltage proportional to -V at the output of circuit 266.

Switching between V and -V with transmission gates 278, 280 and 282, 284 has the effect of multiplying the input by 1 in one switch position and by -1 in the other. This is the same as multiplying by a rectangular time function alternating between values of +1 and -1 with dimensionless units. For the in-phase SIGN square wave which controls gates 278, 280, the function is given by: ##EQU1## When V is multiplied by this quantity, through switching action of gates 278, the following results: ##EQU2## Applying trigonometric identities, this expression may be re-written: ##EQU3## When the ac components (or time-dependent terms) of this expression are removed by the low-pass filter action of filter circuit 288, the only remaining term is: ##EQU4## Thus, a voltage output is provided from circuit 288 which is proportional to the resistive component of the impedance being measured and independent of the reactive component.

Similarly, for the quadrature square wave, which controls gates 282, 284, the function is given by the expression: ##EQU5## Multiplying this by V, applying trigonometric identities, and eliminating ac terms, the following results: ##EQU6## Thus, the voltage output of circuit 290 is proportional to the reactive component of the impedance being measured and independent of the resistive component.

The voltage signals proportional to the real and imaginary impedance components are applied to respective analog-to-digital converters 300, 310 in FIG. 6. Digital values representative of the real and imaginary proportional voltages are output over bus lines D0-D7. Because of the limited bus size, however, the real and imaginary digital values must be separately read by the microcomputer. Accordingly, decoder 320 is provided. Decoder 320 receives a code over address lines A0, A1, and A12-A15. Based upon the address inputs, decoder 320 generates the appropriate signals over lines A00, A01, A02 and A03 to select the appropriate A/D converter for data output to the microcomputer. After selection of converter 300 or 310, the microcomputer outputs the RD signal, which puts converted data on bus 58.

The FLOAT, OUTPUT ENABLE, and range control signals X1, X10 inputs to the circuit in FIG. 5 are generated by portions of the circuit shown in FIG. 6. Beginning with the range control signals X1, X10, those signals are determined by the microcomputer and loaded in latch 330. Also loaded into latch 330 are data bits designated STOP and HOLD. Data is loaded into latch 330 by a strobe signal generated by NOR gate 332. Decoder 340, receiving the same address lines as decoder 320, produces a control output C00, which is one input to NOR gate 332. The other input to gate 332 is the WR signal generated by the microcomputer. Accordingly, when the microcomputer enters commands into latch 330, the microcomputer sets up the data at the latch inputs, outputs the appropriate address to decoder 340, and issues the WR strobe.

The FLOAT and OUT ENABLE signals are generated by circuitry comprising Johnson counter 350 and flip-flop 360. Counter 350 is enabled by the HOLD output from latch 330, and as long as the HOLD output is "high", counter 350 is enabled. Clock pulses for stepping counter 350 through its repeating cycle are derived from the status output of A/D converter 300. A/D converter 300 contains its own clock oscillator, which also provides timing for A/D converter 310. The status output of A/D converter 300 goes high during each conversion and low when conversion is complete. Thus counter 350 counts the number of complete conversions performed by A/D converter 300. As counter 350 is stepped through its output states, the carry output C.sub.o goes "low" beginning with the fifth clock pulse of each repeating cycle, signifying five complete A/D conversions. Inverter 352 inverts the "low" condition to a "high" condition which corresponds to the issuance of OUT ENABLE. This signal causes relay contacts 204, 218 in FIG. 5 to close, connecting the current source circuitry to the measurement terminals. At the next clock pulse, the Q.sub.6 decoded output of counter 350 goes "high" and clocks flip-flop 360. This sets the Q output "high" and issues the FLOAT signal which causes the battery power source in FIG. 5 to be connected to the current source circuitry. Once the current source is both supplied with electrical power and connected to the measurement terminals, measurement current flow is established in the unknown impedance of a circuit under test connected to the measurement terminals. After being clocked twice more (i.e., two more complete data conversions), counter 350 produces a "high" condition on the Q.sub.8 output. This is inverted by inverter 354 to generate the NMI interrupt signal to CPU 52 in FIG. 3. After the microcomputer receives the NMI signal, digital data from A/D converters 300, 310 is read by the microcomputer. When the ninth clock pulse in a cycle is received by counter 350, the Q.sub.9 output issues a reset signal to flip-flop 360, removing the battery power source from the current source circuitry. At the end of the ninth clock pulse, C.sub.o goes "high", disconnecting the current source circuitry from the measurement terminals.

The microcomputer can interrogate as to the status of the A/D converters and the measurement control circuitry and commands. This is accomplished by bus transceiver 370. When the microcomputer interrogates status, decoder 340 is provided with an appropriate address to place a "high" condition on the C01 decode output line. This signal is inverted by inverter 372 and provides an input to OR gate 374. The second input to OR gate 374 is the RD signal from the microcomputer. When the appropriate address is applied to decoder 340 and the RD signal is issued, bus transceiver 370 places the data input information onto bus 58. This information comprises the range selection bits X1, X10, the FLOAT bit from flip-flop 360, and the status outputs from A/D converters 300 and 310.

With reference now to FIG. 7, implementations of the display and keyboard portions of the test set apparatus block diagram in FIG. 2 will be described. Keyboard information and data for display are brought in and out over bus 58. Control over the various portions of the keyboard and display circuitries by the microcomputer is provided through decoder 400, which receives address lines A3, A4 and A12-A15. Decoded output A from decoder 400 enables display driver 402 for operation. Similarly, decoder 400 output B enables display driver 404, and decoder 400 output E enables display driver 406. When decoder 400 output C is selected, it provides a clock signal for latches 408 and 410. Finally, when decoder output F is selected, but transceiver 412 provides address signals on lines A0, A1, and A2 for scanning the keyboard.

Considering first the operation of the display, data for display is provided over lines D0-D5 sequentially to each of the display drivers 402, 404 and 406. The display drivers generate the appropriate output signals to produce several digits of seven segment display readout. Suitably, the displays 22, 24 and 26 are liquid crystal displays and display drivers 402, 404 and 406 are adapted for driving such displays.

In addition to the seven-segment digit display portions, displays 24 and 26 include selectable decimal point displays. Accordingly, circuitry must be provided to receive information from the microcomputer regarding appropriate decimal point location and effect selective turn-on of the appropriate decimal point in each display. Data representing the appropriate decimal points for display on displays 24 and 26 are output by the microcomputer over bus 58 to latch 408. The Q.sub.1 output of latch 408 is applied as one input to Exclusive-NOR gate 414. The other input to gate 414 is a signal from a back plane oscillator included within display driver 404. The output of gate 414 is applied directly to the indicated decimal point in display 24. The output gate 414 is also applied as an input to inverter 416. The output of inverter 416 is applied to the other of the decimal points in display 24.

When the output from Q.sub.1 of latch 408 to gate 414 is a "0", the output of gate 414 will drive the decimal point to which it connects out-of-phase with the back plane oscillator signal. This decimal point is to be driven when the number for display 24 is 10.00-99.99. When Q.sub.1 is a "1", the output of inverter 416 will drive the decimal to which it connects out-of-phase with the back plane oscillator signal. This decimal point is driven when the number for display is 0.000-9.999.

The output of gate 414 is also applied to transmission gate 418. Similarly, the output of inverter 416 is also connected to transmission gate 420. Both transmission gate 418 and 420 are controlled by inverter 422. Input to inverter 422 is from Q.sub.2 of latch 408. This output is a display mode (DSPLY MODE) signal. Inverter 422, when its output is in the low or "0" state, also drives light emitting diode 36, which serves as an indicator that the display mode is in polar form. The Q.sub.2 output of latch 408, when in the low or "0" state, drives directly a light emitting diode 38 which serves to indicate the display as being in rectangular form. The Q.sub.2 output of latch 408 also provides a control signal to transmission gate 430.

When then Q.sub.2 output of latch 408 is "high", and the output of inverter 422 is low, the impedance display mode is in polar form. Transmission gates 418 and 420 are nonconducting, such that the back plane oscillator signal is supplied by resistors 432, 434 to respective ones of the decimal points in display 26. Since the decimal point displays require drive signals which are out-of-phase with the back plane oscillator, those decimal points will not be turned-on.

When Q.sub.2 of latch 408 is "high", transmission gate 436 will be conducting. The back plane oscillator signal, inverted by inverter 440, is then passed through transmission gate 436 to line 444. Accordingly, the display segment driven by line 444 is driven through resistor 442 out-of-phase with the back plane oscillator signal, and will, therefore, be visible.

It, of course, follows that when the Q.sub.2 output of latch 408 is "low", the output of inverter 442 is "high" and transmission gates 418, 420 are conducting. Also, light emitting diode indicator 36 is turned-on to indicate the impedance display as being in rectangular form; and, transmission gate 436 is non-conducting. Thus, depending upon the condition of the Q.sub.1 output of latch 408, gate 414 or inverter 416 will drive the appropriate decimal point in display 26. The decimal point display driven by line 444, however, will not be visible since it is driven through resistor 438 in-phase with the back plane oscillator signal. The remaining output of latch 408, the Q.sub.3 output, drives light emitting diode 48 to indicate that the set is operating in the measurement frequency recall mode.

The remaining light emitting diode indicators on the front panel are driven by the outputs of latch 410. Specifically, the Q.sub.1 -Q.sub.4 outputs of latch 410 drive the Measurement, Key, Auto, Hi, and Lo light emitting diode indicators on the front panel. Latch 410 is loaded with data sent over bus 58 from microcomputer 50. As described previously, the clock signal to latch 410 is derived from the C signal from decoder 400.

In addition to the data for appropriately driving the front panel light emitting diodes, there are also loaded into latch 410 two bits of data which are output from the Q.sub.5 and Q.sub.6 outputs to Exclusive-NOR gates 450, 452. Gate 450 drives the tens of thousands digit in frequency display 22, which either is blanked or displays a 1. Gate 452 drives the minus sign indicator in display 22.

Considering now the keyboard 28 shown in FIG. 7, the keys are connected to a keyboard bus 460 of eight bits, D0-D7, to correspond with bus 58. Data on bus 460 is output through transceiver 462 onto bus 58 when an enable signal is generated by the F output of decoder 400. At the same time bus transceiver 462 is enabled, bus transceiver 412 is also enabled permitting the address on lines A0, A1, and A2 to be brought in to keyboard 28. The microcomputer, while the bus transceivers are enabled, sequentially enables each row of keys in the keyboard. As each row is enabled for output onto bus 460, if any one of the keys is depressed, a digital code representative of the key value is output through bus transceiver 462 onto bus 58 to be read into memory by the microcomputer. After a prescribed period of time during which the keyboard is scanned by the microcomputer, the bus transceivers are disabled.

B. Use and Operation of the Test Set Apparatus

Test set 10 measures the resistive and reactive components of impedance of a circuit under test. The measurement results are displayable in either rectangular or polar form. The test set also calculates for display the return loss of a circuit under test against predetermined standards at a selected measurement frequency.

The frequency range of the signal generator is 20 Hz to 19,999 Hz, selectable in 1 Hz increments. The current source produces at the measurement terminals 50 microamperes of measurement current for high range impedances and 500 microamperes for low range impedances. The measurement circuit detects voltage components of the resulting voltage drop across the terminals which are in-phase and quadrature with the measurement current. In rectangular display form, the impedance ranges are: R: 0 to 1.999 and 0 to 19.99 kilohms, X: 0 to .+-.1.999 and 0 to .+-.19.99 kilohms. In polar display form, the impedance ranges are: Z: 0 to 1.999 and 0 to 19.99 kilohms, .theta.: 0 to .+-.90.degree..

When turned on, test set 10 automatically measures impedance at a measurement frequency of 1004 Hz and displays in rectangular form the phasor impedance being measured at the T and R terminals. If it is desired to measure at a different frequency, for example, 2000 Hz, keystrokes on keyboard 28 of 2-0-0-0 are entered. The frequency display 22 will then read 2000. When the first digit (2) is keyed, impedance measurement stops, with the measurement indicator (LED 44) going out and the key indicator (LED 46) coming on. In the key entry mode, the frequency display 22 shows the key entries. If an error is made in entering the new frequency, the clear key, CLR, may be depressed to permit re-entry of a new value. After the appropriate keystrokes are made for specifying the new measurement frequency, the enter key, ENT, is depressed. This returns the test set apparatus to the measurement mode, and the test set is then measuring and displaying impedance at the new measurement frequency.

A one Hz measurement frequency step change is effected by depression of the Step keys 40 or 42. If key 40 is depressed, the displayed frequency is increased by one Hz. Each time key 40 is depressed, the frequency will be so increased. Thus, after the test set is turned on and 1004 Hz is displayed, if key 40 is depressed four times, frequency display 22 will read 1008 Hz. Key 42 functions in a similar manner to decrease the measurement frequency by one Hz. Depression of key 42 results in a display of 1007 Hz.

The increment or decrement step can be changed to a different size. For a 100 Hz step change, keystrokes of 1-0-0 are made, whereupon frequency display 22 will read 100 Hz. The test set is again in the key entry mode and impedance measurement has stopped. If the ENT key were depressed, the test set would begin measuring impedance at 100 Hz. Instead, by depressing key 40, the frequency display will read 1107 Hz, and the test set will measure impedance at this new measurement frequency. If key 40 is depressed twice more, the measurement frequency will be twice incremented by 100 Hz, and the frequency display will read 1307 Hz. By depressing key 42 three times, the measurement frequency will be decremented three step changes of 100 Hz each, with the frequency display then becoming 1007 Hz. The keyed step change size will return to one Hz if the RCL, STO, or CLR keys are depressed.

If the user wants to measure the impedance of several circuits at different measurement frequencies, the desired measurement frequency values can be stored for recall. For example, suppose it is desired to measure impedance at 404 Hz, 1004 Hz, and 2804 Hz without having to enter them for each measurement. To store a measurement frequency value for 404 Hz, make the following keystrokes: 4-0-4-ENT-STO-1. Similarly, to store a measurement frequency value for 1004 Hz, make the keystrokes: 1-0-0-4-ENT-STO-2. To store the value for 2804 Hz, key in: 2-8-0-4-ENT-STO-3. There is now stored a 404 Hz value at location #1, a 1004 Hz value at location #2, and a 2804 Hz value at location #3. Additional frequency values could be stored at locations #4, 5, 6, 7, 8, 9, and 0. To change a stored measurement frequency value, enter the new value at the desired location.

To recall the measurement frequency value stored in a location for output to the signal generator, depress the RCL key and the numeric key representing the location number. For example, to recall the 404 Hz measurement frequency value in storage location #1, make keystrokes of: RCL-1. When the RCL key is depressed, the test set enters the recall mode in which the depression of a numeric key is interpreted as a recall command to the specified storage register location rather than as the first digit of a new measurement frequency value. The test set immediately begins measuring impedance at the recalled measurement frequency, as indicated by the measurement indicator, LED 44, on the front panel. The recall indicator, LED 48, also stays on, however, indicating that the test set remains in the recall mode. Therefore, to recall the next stored measurement frequency value, the only keystroke required is depression of numeric key 2. The frequency display then reads 1004. Similarly, depression of numeric key 3 results in a display of 2804. To exit the recall mode, the CLR key is depressed, whereupon the frequency display reads 0. Impedance measurement also stops, and the test set enters the key entry mode.

To step through several stored measurement frequency values which have been stored in successive locations, enter the recall mode by keying: RCL-1. Then depress the increment Step key 40. This causes a step to storage register location #2. Another depression of the increment Step key 40 will cause the measurement frequency value in the next higher storage register location, in this case #3, to be recalled. The decrement Step key 42 may be similarly utilized to back through the storage locations.

Test set 10 has two measurement ranges, high (Hi) and low (Lo). At test set turn on, the range selection is in the Auto mode. To change range, depress the SEL key. When the range is changed, the display decimal points move to accommodate the value, thus changing the resolution of the measurement. For example, consider a 600 ohm resistor in series with a 2.2 microfarad capacitor connected to the measurement terminals and a 1004 Hz measurement frequency. In the Lo range, and in the rectangular form display mode, display 24 reads 0.600 kilohms and display 26 reads -0.072 kilohms. If the range is changed to Hi, display 24 reads 0.60 and display 26 reads -0.07.

The test set normally displays impedance in rectangular form, i.e., R and X. If it is preferred to have the impedance displayed in terms of magnitude and phase angle, the display mode key DM is depressed. The 600 ohm resistor in series with the 2.2 microfarad capacitor, at 1004 Hz, has an impedance of 604 ohms at a phase angle of -6.8.degree.. Display 24 then reads 0.604 kilohms and display 26 reads -06.8.degree.. In order to return to display of the rectangular form, the user need only depress the DM key.

Test set 10 can also calculate for display the return loss of a circuit under test at a selected measurement frequency. The calculated value of return loss in dB is displayed on display 26. Return loss is defined as: ##EQU7## where Z is the phasor impedance of a circuit under test and Z.sub.ref is a reference phasor impedance. In test set 10, the microcomputer utilizing a program instruction set calculates return loss in accordance with the above definition for return loss. The phasor impedance Z utilized in the calculation is the measured impedance of a circuit under test connected to the measurement terminals. The microcomputer, of course, calculates the measured phasor impedance by appropriate scaling of the digital values produced by analog-to-digital conversion of the in-phase and quadrature components of the voltage drop across the measurement terminals, which voltages are representative of the resistive and reactive impedance components of the circuit under test. The reference phasor impedance, Z.sub.ref, is either a predetermined value of resistance stored in memory or the impedance of an RC network modeled in the microcomputer and calculated at the measurement frequency by the microcomputer. Furthermore, the RC network, while being of predetermined configuration, may have the values of resistance and capacitance prescribed by the user through keyboard input.

To provide for return loss calculations against a variety of reference phasor impedances, the microcomputer has stored in memory a plurality of keyboard selectable program subroutines. Each of the program subroutines comprises an instruction set serving upon command to instruct predetermined computation processing by the microcomputer CPU to calculate return loss. The program subroutines are as follows:

To run a program subroutine, keys 1 and 3 together are depressed. This puts the test set in the program mode. The frequency display 22 will read P - - - . To enter Program 101, which calculates return loss against 600 ohms in series with 2.16 microfarads, follow with the keystrokes of: 1-0-1. The display 22 will then read P101. Next, depress the ENT key and make a frequency entry, for example 1000 Hz, by making the keystrokes of: 1-0-0-0-ENT. The frequency display will alternate between the frequency and the program number. The calculated return loss in dB is displayed on display 26. Return loss programs 102, 103 and 104 are all entered in the same way as P101.

Program subroutine 105 provides for return loss calculation against the modeled RC network user-keyed component values. All resistance values are entered in ohms and capacitance values in nanofarads. For zero capacitance, an open circuit, depress the ENT key without any value. For infinite capacitance, a short circuit, depress keys 1-9-9-9-9-ENT.

As an example, to measure return loss of a circuit under test against 600 ohms in series with 2.16 microfarads with a shunt capacitor of 0.15 microfarad, make the following keystrokes with the resulting read out in display 22:

The microcomputer in accordance with the instruction set of Program 105 calculates the phasor impedance of the RC network of user selected values at the measurement frequency. This gives a reference phasor impedance value. Then, using the measured phasor impedance of the circuit under test, the microcomputer calculates the return loss.

To check the values of R.sub.1, C.sub.2, and C.sub.3, Program 901 is used. The keystrokes required are as follows: 1-3 together, then 9-0-1-ENT. Display 22 then displays each component value for one second and afterwards returns to executing Program 105.

Program 106 permits calculation of return loss against a precision network, for example, a cable made with the Wilcom T240 artificial cable kit. In making this type of test, the test set first measures the phasor impedance of the precision network at a selected frequency. Then, the test set enters Program 106. To do this, the precision network is connected to the T and R measurement terminals and an impedance measurement is made. With the network still connected, the following keystrokes are made: 1-3 together followed by 1-0-6-ENT. A return loss of 60.0 dB will be displayed, since the network is being tested against itself and 60.0 dB is the maximum value of return loss the test set can display. Next, the network is disconnected and the circuit under test is connected, whereupon the return loss is calculated and displayed.

Test set 10 can further calculate for display weighted-average return loss over prescribed frequency ranges. Also, test set 10 can perform calculations to determine the minimum value of return loss in the frequency range and the frequency at which it occurs.

Return loss may be defined as follows: ##EQU8## For a two wire circuit, return loss can be defined as a function of the circuit phasor impedance and a reference phasor impedance, as noted previously, by the equation: ##EQU9##

By expressing the phasor impedances in rectangular form, the following equation results: ##EQU10##

In making a weighted-average return loss calculation, the return power ration bracketed in the last expression has to be weighted and accumulated. Weighting of the return power ratio for each measurement frequency suitably is in accordance with Bell System Technical Reference PUB-41009, which is hereby incorporated by reference. The weighted return power ratios are summed and stored. To obtain weighted average, the sum is divided by the sum of the weighting factors which were taken at each measurement frequency. The log of the weighted-average return power ratio is then calculated and the result multiplied by -10 to provide the weighted-average return loss over the measurement frequency range.

In determining the minimum value of return loss in a measurement frequency range, unweighted return power ratios are calculated and stored. To determine the minimum return loss, the maximum value of return power ratio is found. For interpolation between return power ratio data points, a parabolic curve-fitting routine describing the general quadratic form y=ax.sup.2 +bx+c is used. By fitting the data points to the curve, an approximate maximum return power ratio value and the approximate frequency at which it occurs can be determined.

To provide for weighted-average return loss calculations, and for determination of the minimum value of return loss, over a prescribed range of frequencies, the microcomputer has stored in memory a plurality of keyboard selectable program subroutines. Each of the program subroutines comprises an instruction set serving upon command to instruct predetermined computation processing by the microcomputer CPU.

If desired, the basic return loss calculation subroutines may be linked with the weighted-average return loss calculation subroutines.

The program subroutines for directing the test set apparatus in making a weighted-average return loss determination on a circuit under test instruct the making of a series of phasor impedance measurements at prescribed, spaced-apart frequencies. The phasor impedances measured are used in calculating return loss impedance ratio against predetermined reference impedance standards. Return loss impedance ratio as used herein is defined as follows: ##EQU11## The inverse square of the return loss impedance ratio for each phasor impedance measurement is determined, which yields the ratio of return power to incident power (i.e., the "return power ratio").

In addition to program subroutines for weighted-average return loss determination, the test set apparatus includes in memory a program subroutine for making unweighted return loss determinations for display. Under the direction of this instruction set, the unweighted return power ratios calculated at the measurement frequencies are summed and averaged for the number of measurements. The result is then multiplied by -10 to yield the average return loss over the frequency range.

Yet another program is provided in the test set apparatus for making weighted and unweighted-average return loss determinations against a precision network, for example, a cable made with the Wilcom T240 artificial cable kit. In making this type of test, the test set first measures the phasor impedance of the precision network at a plurality of measurement frequencies in a frequency range, and stores the measurement values. These stored values are thereafter used as the reference impedance values in the return loss impedance ratios.

The program subroutines described above are designated as follows:

In all the above frequency range sweeps, 2600 Hz will be skipped to avoid exciting the circuit under test at the SF signalling frequency. The data at 2600 Hz will be calculated from an interpolation routine.

Where applicable, the calculated results are displayed sequentially SRL LO, SRL HI, followed by ERL. The data are "identified" in the RESISTANCE/IMPEDANCE display using the following abbreviations:

"SLL" for SRL LO (singing return loss low)

"SLH" for SRL HI (singing return loss high)

"EHO" for ERL (echo return loss)

Abbreviations used in the 505 and 507 programs, respectively, are:

"SPL" for (minimum return loss during a frequency sweep) "LOS" APRL (average power return loss)

The return loss measurements in dB are displayed in the REACTANCE/PHASE ANGLE display in sequence with the "identifying" display entries. The program number is displayed in the FREQUENCY display.

All of the return loss measurements using programs 500-507 are made using a preset reference of 900 ohms in series with 2.16 microfarads. If a different standard such as 600 ohms and 2.16 microfarads is required, program 101 is entered, then one of the "500" series programs is entered. Measurement is now with a reference to a 600 ohm and 2.16 microfarad standard. Any "100" series program or any external network can be used as the reference for return loss measurements. External networks must be stored, first, using program 506. The CLR/EXIT key or the ON/OFF switch will remove the stored information.

In lieu of an external network, the operation of the test set using programs 500-507 can be observed by connecting a resistor between the T and R terminals. 600 ohms nominal is a good value.

To run a program with the present reference, keys 1 and 3 together are depressed. The frequency display 22 will read P - - - . The test is in the program mode. Next enter the program number. For example, for program 500, make keystrokes of: 5-0-0. The display will then read P500. Next, depress the ENT key. The MEAS indicator will come on for several seconds. When the MEAS indicator goes out, the RESISTANCE/IMPEDANCE window will display SLL, SLH and EHO sequentially for a few seconds each. During the same time intervals the REACTANCE/PHASE ANGLE window will display the dB values for these return loss measurements, SRL LO, SRL HI and ERL, respectively. The cycle can be repeated by pressing the ENT key after twice the measure time has lapsed as observed on the MEAS LED. Of course, if the selected program calculates one value, i.e., P503, only SLH (SRL HI) will be calculated and displayed.

Return loss programs 500 through 505, plus 507, are all entered using the above procedure. Programs 506 and 509 are used in conjunction with these programs.

Program 509 is used in place of engaging the ENT key each time you want the measure cycle repeated. It is entered prior to entering the selected return loss program as follows.

Press CLR/EXIT when the MEAS lamp goes out to remove the previous program. To repeat a program continuously, first press keys 1 and 3 together. The FREQUENCY display will read P - - - . Next, make keystrokes of 5-0-9-ENT. Then, enter the return loss program to be repeated. The measure cycle will keep repeating until CLR/EXIT key is depressed.

C. Detailed Program Listing

The following listing is of all the specific steps for carrying out the operational functions and procedures described above. ##SPC1## ##SPC2## ##SPC3## ##SPC4## ##SPC5## ##SPC6## ##SPC7## ##SPC8## ##SPC9## ##SPC10## ##SPC11##

D. Components List

The foregoing description of the invention has been directed to a particular preferred embodiment for purposes of explanation and illustration. It will be apparent, however, to those skilled in this art that many modifications and changes may be made without departing from the scope and spirit of the invention. It is the Applicants' intention in the following claims to cover such equivalent modifications and changes as fall within the scope of the invention as defined by the following claims.