Amplitude Insensitive Voltage-to-Current Converter and Method for Conversion

This invention relates to a unipolar or bipolar amplitude fluctuating voltage signal to constant current signal converter, and more particular to such a converter for providing stable performance in conjunction with current operated devices.

A force balance device in which a current sensitive rebalance component is utilized to close the servo loop is subject to the power transferring inefficiencies inherent in the coupling of a high impedance signal source with a low impedance current sensitive device as well as to the response characteristics of an analog circuit. There is a need therefore for means by which a force rebalance mechanism can be operated utilizing constant current pulses commanded by signal source voltage levels of fluctuating amplitude.

In general, a circuit is disclosed for conversion of an unstable voltage level signal to a stable current level signal which includes circuit means for providing a reference voltage level signal. An operational amplifier having input and output terminals has an impedance coupled to the input with the reference voltage level impressed thereacross. A buffer amplifier is coupled between the input and the output of the operational amplifier and provides a current output scaled by the impedance on the operational amplifier input for a given reference voltage level. A switch is coupled to receive the current output. The switch is actuated by the unstable voltage level signal for passing constant current output pulses having a time duration substantially the same as the time duration of the unstable voltage level signal.

In general it is an object of the present invention to provide electronic circuitry which provides a precise constant current pulse in response to an unregulated voltage level command pulse.

It is another object of the present invention to provide electronic circuitry which directs a precision constant current to a current sensitive device in response to a fluctuating voltage level signal.

It is another object of the present invention to provide electronic circuitry for use with a current sensitive device to obtain uniform current pulses insensitive to the amplitude of actuating signal pulses for use in a digital pulse rebalance system.

It is another object of the present invention to provide electronic circuitry for use with a current sensitive device which provides uniform amplitude current pulses in response to nonuniform amplitude signal pulses having a dwell time and polarity substantially the same as the voltage pulses.

Additional objects and features of the invention will appear from the following description in which the preferred embodiment has been set forth in detail in conjunction with the accompanying drawings.

FIG. 1 is a block diagram of one embodiment of the disclosed invention.

FIG. 2 is an electrical schematic of the embodiment of FIG. 1.

FIG. 3 is a block diagram of another embodiment of the disclosed invention.

FIG. 4 is an electrical schematic of the embodiment of FIG. 3.

FIG. 4a is a partial electrical schematic of another embodiment of FIG. 3.

FIG. 5 is an electrical schematic of one useful form of the disclosed invention.

FIG. 6 is an electrical schematic of a phase splitter for use in the disclosed invention.

FIG. 7 is a timing diagram showing the input and output signals in the circuit of FIG. 6.

FIG. 8 is a timing diagram showing various meter coil current pulse waveforms for the embodiment of FIG. 5.

Referring to FIG. 1 of the drawings, one embodiment of the invention is seen in block form. Means 11 for providing a reference voltage is coupled to a scaling component 12 so that the magnitude of the current I is determined. Scaling component 12 is coupled to the input of an operational amplifier 13 having an output buffer 14 coupled between the output and the input of the operational amplifier 13. A feedback path 16 around operational amplifier 13 includes output buffer 14. Current I flows through feedback path 16. Output buffer 14 produces a current output I.sub.0 which is coupled to a first switch 17 and one terminal of an electric load 18. A second switch 19 is provided which is connected to the opposite terminal of electric load 18. First and second switches 17 and 19 are configured to pass current I.sub.0 when actuated. Actuation control signals are connected to switches 17 and 19 at terminals marked C control and B control respectively.

The block diagram of FIG. 1 may be seen in schematic form in FIG. 2. Zener diode D.sub.1 provides means for producing a reference voltage, which is V.sub.2 -V.sub.1 from FIG. 2. The reference voltage V.sub.2 -V.sub.1 is impressed across an impedance R.sub.1, representing means for scaling 12, which is coupled to the input of operational amplifier 13, signified by component A.sub.1 in FIG. 2. Buffer amplifier 14 is present in the form of the Darlington transistor pair Q.sub.1 and Q.sub.2 coupled between the output and the input of operational amplifier A.sub.1. Buffer amplifier 14 produces an output current I.sub.0 which is directed through the circuit path containing switch 17, depicted by electronic switch Q.sub.3, or the path containing load 18 and switch 19, depicted by the electronic switch Q.sub.4. The path taken by current I.sub.0 is dependent upon voltage control signals presented at terminals B and C which are the B control and C control signals indicated in FIG. 1.

The manner in which the embodiment of FIGS. 1 and 2 operates will now be described. As is known to those skilled in this field of art the characteristics of an operational amplifier include the following: the two input terminals on operational amplifier 13 are always at the same voltage level; there exists a virtual short circuit at the input of operational amplifier 13. The term "virtual" is used to imply that while the voltage between the input terminals is kept at zero, no current actually flows between the input terminals. Furthermore, there is no current flow into or out of either input terminal. The output of the operational amplifier 13 changes to cause the current I in feedback path 16 to assume a value such that the zero voltage difference between input terminals and zero current flow into or out of either input terminal always exists. The Darlington circuit represented by the connection of transistors Q.sub.1 and Q.sub.2 forming a buffer amplifier 14 presents a low input impedance and a high output impedance for preventing interaction between the low impedance load circuit 18 and the relatively high impedance source supplying the potential which is regulated to the value V.sub.2 -V.sub.1. The feedback current I is the emitter current in transistor Q.sub.2, which is substantially the same as the collector current I.sub.0 of transistor Q.sub.2, which is the output current from buffer amplifier 14. The amplitude of the output current I.sub.0 is constant and is determined by the reference voltage V.sub.2 -V.sub.1 and the impedance R.sub.1 across which the reference voltage is impressed.

Voltage signals presented at terminals B and C having unstable amplitude characteristics will actuate transistor switches Q.sub.4 and Q.sub.3 respectively for the dwell time of the voltage level signals at the respective terminals B and C. Constant amplitude output current I.sub.0 is seen to transit transistor switch Q.sub.3 when a voltage level signal sufficient to turn transistor Q.sub.3 "on" is presented to terminal C, for the dwell time of the voltage level signal and independent of the voltage level fluctuations therein. During this dwell time essentially no current will transit load 18 whether transistor switch Q.sub.4 is on or off. Transistor switch Q.sub.4 may therefore be replaced by a conductor, or permanently turned on by an appropriate voltage level signal at terminal B, or alternately turned on and off by inputs to terminal B which are complimentary in time to the inputs to terminal C. When transistor switch Q.sub.3 is in the unactuated or off condition by virtue of an absence of actuating signal at terminal C, and transistor Q.sub.4 is in an "on" condition by virtue of an actuating signal at terminal B, the same constant current I.sub.0 transits load 18 and transistor switch Q.sub.4.

Another embodiment of the present invention is seen in block form in FIG. 3. Like components common to the block diagram of FIG. 1 will be given like item numbers. Means 11 is present for providing a reference voltage level signal which is coupled to each of two parallel paths. The first parallel path includes means 12 for scaling the output current which is connected to the input of an operational amplifier 13. An output buffer 14 is connected between the output and the input of operational amplifier 13 and a feedback path 16 for passing a current I.sub.1 includes an output buffer 14. A current output I.sub.10 is provided at the output of output buffer 14 which is coupled to one terminal of load 18 and to switch 17, which is operated by a fluctuating voltage level signal indicated as C control.

The second parallel path in FIG. 3 includes additional means 21 for scaling an additional constant current output, and an additional operational amplifier 22 having the additional means 21 coupled to the input thereof. An additional output buffer 23 is coupled between the output and the input of operational amplifier 22 and is included in a feedback path 24 around operational amplifier 22. An additional constant current output I.sub.20 is provided at the output of output buffer 23 and is connected to the opposite terminal of load 18 and to one side of an additional switch 26. Additional switch 26 is controlled by a fluctuating amplitude voltage level signal injected as B control in FIG. 3.

The circuit schematic for the block diagram of FIG. 3 is seen in FIG. 4. The circuit for generating constant current I.sub.10 is the same as that described for generating the constant current I.sub.0 in FIG. 2 above. The output of buffer amplifier 14, constant current I.sub.10, is coupled to one terminal of load 18 and to one side of transistor switch Q.sub.3. Transistor switch Q.sub.3 is shown having a base control terminal C for receiving an actuating voltage level signal having an unstable amplitude. The circuit for providing constant current I.sub.20 is the same as that described before for generating constant current I.sub.0 in FIG. 2, wherein reference voltage V.sub.2 -V.sub.1 is impressed across an impedance R.sub.2 representing additional means 21 for scaling the constant current I.sub.20. Impedance R.sub.2 is coupled to the input of operational amplifier A.sub.2 representing block 22 in FIG. 3. The Darlington transistor pair Q.sub.5 and Q.sub.6 represents the output buffer 23 and provides an output current I.sub.20 which has a constant amplitude determined by the reference voltage V.sub.2 -V.sub.1 and the impedance R.sub.2.

As described above the emitter current I.sub.2 for transistor Q.sub.6 is substantially the same as the constant output current I.sub.20 from output buffer 23. The output current I.sub.20 is coupled to the other terminal of load 18 and to transistor switch Q.sub.7 representing switch 26 of FIG. 3. Transistor Q.sub.7 has a terminal B to which is applied actuating voltage level signals having unstable amplitude characteristics. The manner in which the embodiment of FIG. 4 operates is described as follows. The generation of constant output currents I.sub.10 and I.sub.20 is the same as that described for the generation of constant current output I.sub.0 in FIG. 2 above. Amplitude fluctuating voltage level signals are alternately applied at terminals B and C of FIG. 4, thereby alternately directing constant current levels I.sub.10 and I.sub.20 through load 18 respectively. Current I.sub.10 will transit load 18 for the dwell time of the amplitude unstable voltage level applied to terminal B. No current flows through transistor switch Q.sub.3 as there is no signal at terminal C and a current through transistor switch Q.sub.7 is the sum of the currents I.sub.10 and I.sub.20. Upon removal of the unstable voltage level signal from terminal B and replacement of such a signal on terminal C, transistor switch Q.sub.7 is opened, no current flows therethrough, and current I.sub.20 flows in a direction opposite to that of current I.sub.10 through load 18 for the dwell time of the voltage level signal on terminal C. Transistor switch Q.sub.3, being in the conducting condition, passes the sum current I.sub.10 and I.sub.20. In this fashion, constant current having alternate polarity is passed through load 18.

Turning now to FIG. 5, one use of the disclosed invention is shown. FIG. 5 is similar to FIG. 4 in many respects, and like items have like item numbers and designations. The circuit structure and manner in which constant output currents I.sub.10 and I.sub.20 are obtained is described above. The electrical load is a meter coil 27 in this embodiment. Meter coil 27 is a current actuated device providing for angular displacement of the coil as a function of current therethrough when rotationally suspended in a magnetic field. Meter coil 27 is mechanically coupled to an indicator needle 28 for motion over a dial face 29.

A phase splitter 31 is shown receiving an input signal and providing two outputs through diodes D2 and D3 to terminals B and C respectively. Reference is made to FIG. 6 for the structural details of phase splitter 31 and to FIG. 7 for a timing diagram showing the time relationships between the input signal and the output signals coupled to terminals B and C. In this configuration the input signal is connected to the primary winding 32 on a transformer having the polarity indicated. A pair of secondary windings 33 and 34, having polarity indicated in FIG. 6, are coupled to terminals B and C through diodes D2 and D3 respectively. The input/output signal relationships seen in FIG. 7 result. An alternating input signal having the arbitrary amplitude envelope seen at 36 induces an in phase signal in the secondary winding 33 having the amplitude envelope 37 which is presented at terminal B. Output 37 at terminal B may be seen to have a dwell time which is substantially t2-t1. An amplitude envelope 38 is seen for the AC signal induced in secondary winding 34 and presented through diode D3 to terminal C. The dwell time of the fluctuating voltage level signal at terminal C is seen to be t3-t2, and the signal is out of phase with the input signal. In this fashion, reversing polarity control or drive signals are presented to the actuating terminals for the transistor switches Q.sub.3 and Q.sub.7 as like polarity signals for reversing the direction of the constant current through the meter coil 27 in accordance with the polarity of the control signal. Constant current levels are therefore directed to meter coil 27 in accordance with the polarity of the input voltage level signal and without regard for the amplitude fluctuations in the input voltage level signal.

It should be noted that transistor switches Q.sub.3 and Q.sub.7 could be replaced by any other type of suitable switch, such as field effect transistors with gates connected to terminals B and C. Any device displaying nearly a short circuit in the presence of a control signal and nearly an open circuit in the absence of a control signal would be a suitable switch FIG. 4a shows a partial schematic similar to that of FIG. 4 which is suitable. It is understood that field effect devices are also usable in the embodiment of FIG. 2 in place of bipolar transistors Q3 and Q4 shown.

Since the amplitude of the constant current outputs I.sub.10 and I.sub.20 are determined by the magnitudes of the impedances R1 and R2, the constant current outputs may be scaled by adjustment of the R1 and R2 magnitudes. Thus, the scaling of the deflection of needle 28 on dial face 29 may be adjusted. Moreover, a current sensitive component such as meter coil 27, may exhibit a bias. The bias referred to is defined as a characteristic whereby the meter coil deflects through a greater angle for one polarity of current compared to the opposite polarity of current having the same magnitude or that produced by a mechanical spring set off zero. In such an event the impedances R1 and R2 may be adjusted singly to remove the bias in the current sensitive component.

FIG. 8 shows the manner in which the apparatus of FIG. 5 displays an indication on dial face 29. Since the fluctuating voltage level signals at B and C are time coincident with the constant currents I.sub.20 and I.sub.10 respectively through metered coil 27, the timing diagram of FIG. 8 is expressed in terms of the constant currents. In the upper diagram of FIG. 8, equal time duration signals are presented to terminals B and C of FIG. 5 providing for alternating pulses of I.sub.20 and I.sub.10 respectively through meter coil 27 of equal dwell periods, T.sub.1 and T.sub.2. As indicated to the right of the upper diagram, indicator needle 28 will indicate a zero reading on dial face 29. In the middle of diagram of FIG. 8 an unstable voltage level signal is presented to terminal C of FIG. 5 continuously. Consequently, as indicated to the right of the middle diagram indicator needle 28 is deflected full scale on dial face 29. In the lower diagram of FIG. 8 an actuating voltage level signal is presented at terminal C in FIG. 5 for the period T.sub.1 and the terminal B for the period T.sub.2. Since transistor switch Q.sub.7 directs constant current I.sub.10 through meter coil 27 for three quarters of the time total of periods T.sub.1 and T.sub.2, indicator needle 28 will take a position halfway between zero and the positive extreme seen on dial face 29, or three quarters of the full deflection of needle 28.

The device disclosed herein is particularly useful for current pulse rebalance applications where a digital input is available to terminals B and C. The invention disclosed herein may be used in a unipolar configuration such as that disclosed in conjunction with FIG. 2 or in a bipolar configuration such as that disclosed in conjunction with FIG. 4. Conversion of voltage pulses to time coicident constant current pulses is obtained, wherein the current pulses are insensitive to voltage pulse amplitude fluctuation. Wide voltage amplitude swings can be applied at the load terminals without load current amplitude change.