Electronic Monitoring System With Low Energy Consumption in Quiescent State

This application discloses subject matter described and claimed in my copending applications Ser. No. 782,400, filed 29 Mar. 1977, and Ser. No. 786,805, filed 12 Apr. 1977.

My present invention relates to an electronic monitoring system, as used for example in a proximity sensor, wherein a preferably contactless detector, sensitive to an ambient condition, generates an output signal which varies with a change in that condition, e.g. with the approach of a metallic element, to increase (or decrease) the current flow through a load designed to indicated such change.

Such a monitoring system has been described in my prior U.S. Pat. Nos. 3,737,010, 3,747,011, 3,747,012, 3,919,661, 3,932,774, 3,932,803 and 3,935,542. As particularly disclosed in U.S. Pat. No. 3,932,803, the contactless motion detector may comprise an oscillator and an associated trigger amplifier whose energizing circuit includes a storage capacitor which is charged via a two-conductor supply network from a source of pulsating direct current by way of a constant-current unit in parallel with the anode/cathode path of an output thyristor acting as a voltage-controlled variable-impedance means. In one embodiment of the system described and claimed in the latter patent, the output thyristor and the constant-current device lie in series with an ancillary thyristor which shunts the storage capacitor and is separated therefrom by a decoupling diode. When the output thyristor is triggered by a signal from the oscillator, the ancillary thyristor is fired through a Zener diode inserted between its anode and its gate; thus, the two series-connected thyristors conduct simultaneously in series with a load such as a relay whose operated (or unoperated) state indicates an abnormal condition, e.g. the fact that the oscillator has detected the approach of a metallic element.

Though the system just described operates generally satisfactorily, a drawback resides in the fact that the load is energized through a resistance representing the sum of the resistance of the two series-connected thyristors in their conductive state. To eliminate this drawback, I have disclosed in my first-mentioned copending application Ser. No. 782,400 an improved system in which the resistance in series with the load is still further reduced in its high-current phase to increase the sensitivity of the system.

According to that improvement there is provided a first electronic control device having a pair of main electrodes respectively tied to the leads of the supply network and further having an input electrode, or gate, connected to one of its main electrodes (i.e. its anode if that control device is a thyristor or similar binary electronic switch such as a triac) through an operating circuit which includes a second electronic control device with input connections to the detector for changing the conductivity of the first control device in response to variations in the output signal of that detector, thereby modifying the energization of the load. Thus, the current flow through the load in a state of high conductivity of the first control device is limited practically exclusively by the resistance of that one device (aside from the internal resistance of the current source which preferably, as in the systems of my prior patents, includes a full-wave rectifier). Instead of binary electronic switches such as thyristors I may also use analog-type current regulators such as transistors establishing an entire gamut of energization rates in lieu of just a switchover between "on" and "off" states. Furthermore, the active component of the detector need not be an oscillator but could be an impedance bridge, a field plate or any of a variety of electromagnetic, photoelectric or other transducers responding to a predetermined change of an external condition to be monitored.

The storage capacitor insuring the continued energization of the detector, regardless of the state of conductivity of the electronic control devices, is connected across the leads of the supply network in series with a current-limiting device, more specifically a constant-current unit as in my prior U.S. Pat. No. 3,932,803. A decoupling diode is inserted between the storage capacitor and the ancillary thyristor (or other electronic switch constituting the aforementioned second control device), triggerable by the detector, activating the main thyristor (or other electronic switch representing the first control device) whose conduction raises the load current to its maximum value. The operating circuit lying between the anode and the gate of the main thyristor advantageously includes an impedance, preferably an electronic breakdown device such as a Zener diode, delaying the firing of the main thyristor after the ancillary thyristor has been triggered. During this brief delay period, the conduction of the ancillary thyristor recharges the storage capacitor through the decoupling diode while the constant-current unit is practically short-circuited by that conduction. A stabilizing resistor inserted between the gate and the cathode of the main thyristor lies in series with the aforedescribed operating circuit to limit the flow of gate current in that thyristor. The firing of the main thyristor virtually short-circuits the series combination of ancillary thyristor, Zener diode and stabilizing resistor so that the latter thyristor ceases to conduct. With pulsating or raw-rectified current supplied by the source, the main thyristor is also cut off at the end of each pulsation during which it has been rendered conductive; this results in a periodic recharging of the storage capacitor at the beginning of each new half-cycle (or full cycle in the case of half-wave rectification).

In the system just referred to, as well as in those of my prior patents wherein a storage capacitor shunted by a charge-limiting electronic breakdown device such as a Zener diode is charged through a constant-current unit, the breakdown device draws practically the full output current of the constant-current unit upon becoming conductive when the storage capacitor reaches a predetermined threshold voltage. This current flow is generally considerably greater than would be necessary in order to maintain conduction of the breakdown device until the capacitor charge has decayed below the sustaining level. It therefore constitutes a wasteful expenditure of energy and may also lead to objectionable heating of the load which lies effectively in series with the breakdown device. In a binary switching system, in which the load is to be in a state of minimum energization when the main or output thyristor is cut off, the existence of a large load current in the quiescent state is especially undesirable.

My other copending application referred to above, Ser. No. 786,805, describes and claims a further improvement through which the sustaining current drawn by a conductive charge-limiting electronic breakdown device is minimized. According to this latter improvement, a system of the general type described and claimed in my prior U.S. Pat. No. 3,932,803 is modified by connecting an electronic breakdown device such as a Zener diode between one of the output leads of the direct-current source, i.e. a lead which has an extension included in the energizing circuit for the motion detector, and a control input of a constant-current unit normally supplying charging current to a storage capacitor in series therewith, this Zener diode being nonconductive as long as the charging voltage of the capacitor lies below a predetermined level. When that level is reached, the voltage across the Zener diode equals its breakdown threshold whereupon that diode conducts and establishes a reference potential which, together with the rising capacitor voltage, throttles -- possibly down to complete cutoff -- the flow of charging current into the capacitor while drawing a reduced sustaining current from a section of the constant-current unit which lies between that control unit and the other lead of the d-c source.

More particularly, and as also disclosed in my prior U.S. Pat. No. 3,932,803, a constant-current unit for the charging of a storage capacitor in the presence of widely varying supply voltages (ranging, for example, between 40 and 250 V) comprises two parallel impedance branches, one of these branches including a transistor serving as an analog-type current regulator while the other branch comprises a voltage divider with a tap tied to the base of this current-regulating transistor. In the improved system, this tap constitutes the aforementioned control input of the constant-current unit whereby the sustaining current for the charge-limiting Zener diode is drawn only through one or more resistors forming part of the voltage divider; with the emitter of this transistor connected to the storage capacitor (usually through a resistance), this emitter-follower connection progressively reduces the conductivity of the transistor once the Zener threshold has been exceeded.

Even with the improvements first disclosed in my aforementioned copending applications, however, the load in its quiescent state is still traversed by a significant current drawn by a pilot transistor, or equivalent electronic switch, which is cut off by the oscillator only when the variable-impedance means in series with the source and the load is to be made highly conductive by the triggering of a thyristor of the like forming part thereof. In my prior systems, including that of U.S. Pat. No. 3,932,803, this pilot transistor is shown connected across the two supply leads in series with a resistor which limits but obviously does not cut off the current flow therethrough in the quiescent state.

Thus, the object of my present invention is to provide a further improvement in such an electronic monitoring system which eliminates this parasitic current flow in the state of minimum energization of the load.

I realize this object, in accordance with my present invention, by the provision of electronic switch means, specifically a pilot transistor, connected between an input electrode (gate) of a thyristor or other electronic control device and a supply lead which does not form part of the energizing circuit for the detector, i.e. the lead -- hereinafter assumed to be positive -- which is connected to the storage capacitor via vurrent-limiting means such as a high-ohmic resistor or, preferably, a constant-current unit. The input circuitry of the pilot transistor or equivalent electronic switch means, connected to the detector, cuts off that switch means under conditions of minimum energization of the load, thus when the output thyristor or other control device in series with the load is in its high-resistance state.

According to a more particular feature of my invention, the connection between the positive supply lead and the pilot transistor includes a portion of the current-limiting means forming part of the voltage-generating network through which the storage capacitor of this network is charged. If the current-limiting means is a constant-current unit, as in the system of my last-filed copending application, that portion advantageously comprises a current-regulating transistor in a branch of this unit whose base is tied to a tap on a voltage divider constituting another branch thereof; that tap is preferably connected to the opposite (negative) supply lead through an electronic breakdown device such as a Zener diode for limiting the charging current as described in that copending application, the current-regulating transistor acting as an emitter-follower.

Pursuant to a further advantageous feature of my invention, a protective transistor complementary to the pilot transistor is connected in cascade with the latter transistor, and with the current-regulating transistor of the constant-current unit, between the input electrode (gate) of the electronic control device and the opposite (negative) supply lead for positively preventing untimely conduction of that control device. For this purpose the operating circuit for the control device drives the two complementary transistors via inversion means so as to cut off the pilot transistor whenever the protective transistor conducts, and vice versa.

The above and other features of my invention will now be described in detail with reference to the accompanying drawing in which:

FIG. 1 is a block diagram of an electronic monitoring system according to my invention;

FIG. 2 is a more detailed circuit diagram of the system of FIG. 1 embodying the present improvement; and

FIG. 3 is a circuit diagram showing a partial modification of the system of FIG. 2.

FIG. 1 shows the basic components of a monitoring system embodying my invention, namely an oscillator 6 and a trigger amplifier 9 together constituting a detector, a current-responsive load 5 such as a relay in a supply circuit whose two conductors 2, 3 are connected via respective bus bars 7, 8 across an a-c source 4, a full-wave rectifier 14 inserted between the supply conductors 2, 3 and a pair of leads 18, 21 extending to a variable-impedance network 12' whose impedance controls the load current in response to signals from oscillator 6, and a voltage-generating network 13 also receiving the output of rectifier 14 by way of leads 18 and 21. Network 13 is connected to amplifier 9 by leads 10, 11, 18' and 21'. Leads 11 and 18' control a protective transistor 35 (FIG. 2) within network 13 whereas leads 10 and 21' are part of an energizing circuit supplying the detector stages 6 and 9 with operating voltage; supply leads 18' and 21' are extensions of conductors 18 and 21 which are the positive and negative output leads of d-c source 4, 14. Components 6, 9, 12', 13 and 14 form part of a proximity sensor 1 of the general type described in my prior U.S. patents, particularly U.S. Pat. No. 3,932,803 whose disclosure is hereby incorporated by reference into the present application. Negative leads 21, 21' are shown to be grounded.

In FIG. 2 I have shown only part of component 9 along with components 12' and 13 of the system of FIG. 1 since the rest conforms to the illustration of U.S. Pat. No. 3,932,803. Component 12' comprises an ancillary thyristor 12 with an anode tied to positive supply lead 18 carrying pulsating voltage +V, a gate connected to a lead 16 and a cathode provided with a lead 23. Component 13 includes a PNP pilot transistor 32 in cascade with protective NPN transistor 35; the collectors of these complementary transistors 32, 35 are connected to gate lead 16. Transistor 35 has its emitter tied directly to negative supply lead 21 while its base is joined to lead 11 which is normally driven positive by oscillator 6, via a voltage divider 9e, 9f as also shown in U.S. Pat. No. 3,932,803, so that gate lead 16 is practically at voltage -V, maintaining the thyristor 12 in its nonconductive state.

The connection between the base of transistor 35 and the junction of resistors 9e, 9f within amplifier stage 9 includes an NPN transistor 36 whose base is joined to that junction and whose emitter is tied, in a Darlington circuit, to the base of another NPN transistor 37 with grounded emitter, the collector of the latter transistor being connected through a resistor 38 to the base of pilot transistor 32. The collector of transistor 36 is connected to lead 11 via a stack of diodes 39 and to positive lead 18' by way of a resistor 40; diodes 39 introduce a voltage drop sufficient to cut off the protective transistor 35 when transistors 36, 37 and 32 saturate. Thus, the Darlington pair 36, 37 constitutes an inverter driving the two complementary transistors 32, 35 in opposite senses.

A storage capacitor 29, inserted between leads 10 and 21, supplies operating current to components 6 and 9. Capacitor 29 lies in series with a constant-current unit 15 of the type described in U.S. Pat. No. 3,932,803, connected to supply lead 18. The junction J of current limiter 15 and capacitor 29 is connected through a decoupling diode 25 to the cathode lead 23 of thyristor 12.

A main thyristor 22, also forming part of component 12', shunts the series combination of current limiter 15 and capacitor 29 by having its anode tied to lead 18 and its cathode joined to lead 21; a gate lead 24 of thyristor 22 is connected to lead 23 through a Zener diode 19 and to lead 21 (ground) through a stabilizing resistor 26.

In accordance with the teachings of my last-filed copending application Ser. No. 786,805 referred to above, the charging voltage of capacitor 29 is limited by a Zener diode 27 connected between supply lead 21 and a control input 30 of constant-current unit 15. This unit includes a current-regulating transistor 41 of NPN type having its collector tied to positive supply lead 18 by way of a resistor 73 and having its emitter connected to junction J through a resistor 46 in series with a stack of diodes 34 establishing a substantially constant voltage drop thereacross. This branch is shunted by a voltage divider comprising a resistor 43 in series with a Zener diode 44. The base of transistor 41 is connected to a tap of the voltage divider represented by the junction of impedance elements 43 and 44, the potential of this junction being held constant with reference to junction J by the Zener diode 44 as long as the charge-limiting Zener diode 27 is nonconductive. Since the voltage drop across the base/emitter path of transistor 41 is also substantially constant, the current flowing through transistor 41 is essentially stable provided that the voltage on lead 18 is high enough (e.g. upward of 40V) to saturate that transistor. The emitter of NPN transistor 41 is also connected via a lead 33 directly to the emitter of PNP transistor 32; thus, transistor 41 lies in cascade with transistor 32 and 35.

As long as detector 6, 9 does not have a large enough output current to turn on the Darlington pair 36, 37, high voltage on the collector lead of transistor 36 saturates the transistor 35 while the blocking of transistor 37 cuts off the transistor 32. Under these circumstances, thyristor 12 does not fire and the load current remains low.

When Zener diode 27 breaks down, input lead 30 is held at a fixed potential (e.g. of about 6V) with reference to that of leads 21, 21' so that the voltage difference between the base and the emitter of transistor 41, which is biased through diodes 34 and resistor 46 by the charging voltage of capacitor 29, progressively diminishes as long as the current flowing into the capacitor exceeds the current drawn by components 6 and 9. This also cuts off the Zener diode 44 whereby unit 15 passes only a residual charging current besides the small sustaining current for Zener diode 27 traversing the resistor 43.

When transistors 36, 37 become conductive, transistor 35 is cut off as the voltage drop across diodes 39 substantially matches the combined drop on the inputs of the Darlington pair. Near-ground potential on the base of transistor 32 also lowers the emitter potential of transistors 32 and 41 to less than the threshold voltage of Zener diode 27 so that enough current flows through resistor 73 and transistors 41, 32 to trigger the thyristor 12.

In FIG. 3 I have shown another constant-current unit 15' wherein the emitter resistor 46 of transistor 41 is connected across the base 50 and the emitter of an ancillary transistor 52 replacing the Zener diode 44 of the voltage divider of FIG. 3; the collector of transistor 52 is connected to resistor 43 through another resistor 68. Furthermore, a transistorized shunt circuit 49 is connected across part of the voltage divider 43, 68, 52, this shunt circuit including a supply connection 48 tied to the base 42 of transistor 41, an input connection 53 extending from the collector of transistor 52, and an output connection 55 extending to a junction K on the emitter side of the resistor 46 and to the base 50 of transistor 52.

A damping capacitor 69 bridges the collector and the base of transistor 52. Transistor 41 is provided with a protective collector resistor 73 and has its emitter connected to resistor 46 through a threshold-responsive diode 70 represented by an NPN transistor having its base returned to its collector.

Shunt circuit 49 comprises two complementary transistors in tandem, namely a PNP transistor 60 and an NPN transistor 61. Transistor 60 has an emitter tied to lead 48, a base 63 tied to lead 53 and a collector tied to the base 66 of transistor 61 whose collector is joined to lead 48 and whose emitter is joined to lead 55.

As further shown in FIG. 3, the control input 30 of constant-current unit 15' is again tied to the base 42 of transistor 41 and also to a tap of the voltage divider here represented by branch 43, 68, 52. The emitter of transistor 41 is joined via lead 33 to the emitter of pilot transistor 32 (FIG. 2). The operation of this unit in the nonconductive state of Zener diode 27 has been fully described in my prior U.S. Pat. No. 3,932,803. When Zener 27 breaks down, the voltage of base 42 of transistor 41 is again stabilized to throttle the flow of charging current through branch 73, 41, 70, 46.

It will be understood that the threshold voltage of Zener diode 27 should exceed the desired maximum voltage of storage capacitor 29 by the substantially constant voltage drop which normally (i.e. in the nonconductive state of that Zener) exists between control input 30 and junction J.

As long as thyristor 12 (FIG. 2) is cut off, thyristor 22 also remains nonconductive. During each half-cycle of a-c source 4 (FIG. 1), enough current passes the unit 15 or 15' to charge the capacitor 29 to the breakdown voltage of Zener diode 27 in order to keep oscillator 6 and amplifier 9 operating. When the oscillator detects the approach of a metallic element, lead 11 is driven negative and triggers the thyristor 12 as described above. The current now traversing this thyristor, passed by the diode 25, replenishes the charge of capacitor 29 at the beginning of the next half-cycle and, before Zener diode 27 conducts, breaks down the Zener diode 19 whose threshold is lower than that of diode 27. This action ignites the thyristor 22 which now short-circuits the remainder of the supply network shown in FIG. 2 and causes the flow of a large load current through the relay 5 (FIG. 1) which operates to indicate the response of detector 6, 9. Aside from rectifier 14 and the internal resistance of source 4, only the small resistance of the conductive thyristor 22 lies at this point in series with the load.

Relay 5 may be sufficiently slow-releasing to remain operated between current pulses when thyristors 12 and 22 are both cut off. If the detected off-normal condition persists, the thyristors are sequentially retriggered at the beginning of the next half-cycle with brief recharging of capacitor 29 through diode 25 just before the firing of thyristor 22.