QUADRATURE MODULATION TRANSCEIVER AND PARAMETER ESTIMATING METHOD FOR IQ IMBALANCE CALIBRATION

[0001] 1. Field of the Invention

[0002] The present invention relates to a communication apparatus and calibration method thereof, and more particularly, to a quadrature modulation transceiver and parameter estimating method for calibrating IQ imbalance.

[0003] 2. Description of the Prior Art

[0004] Please refer to FIG. 1. FIG. 1 is a diagram of a prior art quadrature amplitude modulation (QAM) transceiver 100. As shown in FIG. 1, the transmitting end of a prior art QAM transceiver 100 utilizes mixers 110 and 120 to up-convert an in-phase component I_t and a quadrature-phase component Q_t of a base-band signal or an intermediate-frequency (IF) signal, where an in-phase transmitting carrier S_It and a quadrature-phase transmitting carrier S_Qt are respectively mixed with the in-phase component I_t and the quadrature-phase component Q_t to generate a transmitting signal S_t for transmission by an antenna 150. At the receiving end of a prior art QAM transceiver 100, a receiving signal S_r is received by an antenna 160 and fed into mixers 130 and 140. Mixers 130 and 140 respectively utilize an in-phase receiving carrier S_Ir and a quadrature-phase receiving carrier S_Qr to down-convert the receiving signal S_r and generate an in-phase component I_r and a quadrature-phase component Q_r. Ideally, the in-phase transmitting carrier S_It and the quadrature-phase transmitting carrier S_Qt are mutually orthogonal and their amplitudes are identical. Similarly the in-phase receiving carrier S_Ir and the quadrature-phase receiving carrier S_Qr are ideally mutually orthogonal with their amplitudes being identical. This will result in a balanced IQ in the QAM transceiver 100. However, under practical conditions, IQ imbalance is usually inherent in the QAM transceiver—the key reason of IQ imbalance being that the lengths of traces designed on the circuit layout are not ideally matched. Additionally, IQ imbalance is largely affected by increasingly higher carrier frequencies even though the lengths of traces are just slightly different.

[0005] There are four main error parameters commonly used in IQ imbalance modeling. These parameters are: the phase error of the transmitting carrier, the amplitude error of the transmitting carrier, the phase error of the receiving carrier, and the amplitude error of the receiving carrier. Many prior art methods for calibrating IQ imbalance are proposed, most of which additionally utilizing a sequential searching method. Through sequential searching, each error parameter is adjusted one at a time, with subsequent adjustments to other parameters following to fine tune the IQ until calibration is systematically completed. The error parameters corresponding to the transmitting end are usually adjusted first, and then the error parameters corresponding to the receiving end are adjusted. However, a cost function is still necessary for measuring the current level of IQ imbalance to act as a reference in adjusting the error parameters. The sequential searching method is finished when the value of the cost function reaches a local minimum. These methods for calibrating IQ imbalance using simple concepts can be implemented easily, however many errors can still result.

[0006] Firstly, the transmitting end and the receiving end are not calibrated in the same procedure. Alternatively, the error parameters corresponding to the transmitting end are first calibrated followed by the error parameters corresponding to the receiving end. This method of calibration leads to error propagation if the calibration is incorrect. More specifically, a potential calibrating error at the transmitting end would be propagated to the receiving end, further increasing the resulting calibration error at the receiving end. Secondly, according to these conventional methods, extra analog circuits are necessary to perform follow-up signal processing and then extra error items are resulted while the calibration procedure is performed. This will greatly influence and affect the accuracy of the calibration procedure. Finally, the use of a cost function and searching algorithm does not depend only on minimized error parameters. In performing the calibration procedure for IQ imbalance, the convergence and convergence speed of the cost function are considered largely, and the convergence and convergence speed of the cost function are influenced by the cost function itself, the searching algorithm, and the initial state. The choice of a cost function shall not be overly complicated such that the acceptable convergence speed will not be too slow. Since the error parameters are mutually dependent, an optimum searching method should be a two-dimension searching method. However, the complexity involved with a two-dimension searching method is much higher and difficult to arrive at a convergence. This is the main reason why a sequential searching method is typically adopted according to the prior art, although a lower degree of accuracy is attained. All of these in the prior art are difficult and deficient.

[0007] It is therefore one of the objectives of the claimed invention to provide a loopback component to generate a loopback signal. The loopback signal possesses a different amplitude or different phase in order to calibrate IQ imbalance in the quadrature modulation transceiver (for example, the QAM transceiver). In this way, a parameter estimating method is provided to solve the above-mentioned problems.

[0008] According to the present invention, a quadrature modulation transceiver is disclosed. The quadrature modulation transceiver comprises a transmitter, a loopback component, a receiver, and a calibration unit. The transmitter is utilized to receive an in-phase component and a quadrature-phase component corresponding to an input signal, and generating a transmitting signal by up-converting the in-phase component and the quadrature-phase component according to an in-phase transmitting carrier and a quadrature-phase transmitting carrier respectively. The loopback component is coupled to the transmitter and is utilized to provide a loopback parameter to adjust the transmitting signal such that a loopback signal is generated. The loopback component also comprises a plurality of sets of loopback parameters, with the loopback parameter being one set of the plurality of sets of loopback parameters. The receiver is coupled to the loopback component and is utilized to down-convert the loopback signal to generate an in-phase component and a quadrature-phase component corresponding to a receiving signal according to an in-phase receiving carrier and a quadrature-phase receiving carrier respectively. The calibration unit is coupled to the transmitter and the receiver and is utilized to generate the input signal, to receive the receiving signal, and to compute calibration parameters for IQ imbalance of the quadrature modulation transceiver.

[0009] According to the present invention, a method for estimating calibration parameters for IQ imbalance applicable to a quadrature modulation transceiver is disclosed. First, an in-phase component and a quadrature-phase component is received from an input signal and up-converted to generate a transmitting signal according to an in-phase transmitting carrier and a quadrature-phase transmitting carrier respectively. Second, the method utilizes a loopback parameter to adjust the transmitting signal such that a loopback signal is generated. Third, the method down-converts the loopback signal to generate an in-phase component and a quadrature-phase component of a receiving signal according to an in-phase receiving carrier and a quadrature-phase receiving carrier respectively. Finally, the method computes the calibration parameters of IQ imbalance of the quadrature modulation transceiver.

[0010] These and other objectives of the present invention will no doubt become obvious to those of ordinary skill in the art after reading the following detailed description of the preferred embodiment that is illustrated in the various figures and drawings.

[0017] Please refer to FIG. 2. FIG. 2 is a diagram of an embodiment of the QAM transceiver 200 according to the present invention. As shown in FIG. 2, the QAM transceiver 200 comprises a calibration unit 205, a plurality of mixers 210, 220, 230, 240, a plurality of path selectors 250, 260, a loopback component 270, and a plurality of antennas 280, 290. When the QAM transceiver 200 is operating, the mixer 210 receives an in-phase component I_t of a received signal and up-converts the in-phase component I_t with an in-phase transmitting carrier S_It. Another mixer 220 receives the quadrature-phase component Q_t of the received signal and up-converts the quadrature-phase component Q_t with a quadrature-phase transmitting carrier S_Qt. The output signals from mixers 210 and 220 are then combined to generate a transmitting signal S_t. Under normal transceiver conditions, the path selector 250 selects and transmits the transmitting signal S_t to the antenna 280 for broadcast.

[0018] As to the receiving end, an antenna receiving signal S_a is received by the antenna 290, and then transmitted to the path selector 260. The path selector 260 then selects and outputs the receiving signal S_r into mixers 230 and 240 respectively. The mixer 230 down-converts the receiving signal S_r with an in-phase receiving carrier S_Ir to generate an in-phase component I_r corresponding to an output signal, while the mixer 240 down-converts the receiving signal S_r with a quadrature-phase receiving carrier S_Qr to generate a quadrature-phase component Q_r corresponding to the output signal. Please note that under normal transceiver conditions, the receiving signal S_r is identical to the antenna receiving signal S_a, so that the path selector 260 simply bypasses the antenna receiving signal S_a. Assuming an ideal operating condition, the phase difference between an in-phase transmitting carrier S_It and a quadrature-phase transmitting carrier S_Qt is 90°, with their amplitudes are identical. The in-phase component I_t and the quadrature-phase component Q_t are both sinusoidal signals with frequency f₂, which are represented as follows:

I_t=a×cos(2πf₂t)   Eq. (1)

Q_t=a×sin(2πf₂t)   Eq. (2)

[0019] In the above equations, ‘a’ represents the amplitude of the respective signal, while ‘t’ represents time.

[0020] If the in-phase carrier S_Ir and the quadrature-phase carrier S_Qr are sinusoidal signals with a high frequency f₁ and represented as cos(2πf₁t) and −sin(2πf₁t) respectively, the in-phase component I_r and the quadrature-phase component Q_r are mixed and combined to generate a transmitting signal S_t through the in-phase carrier S_Ir and the quadrature-phase carrier S_Qr respectively. The transmitting signal S_t can be represented with a summation of the following equations:

$\begin{array}{cc}a\times \cos \left(2\pi f_1t\right)\cos \left(2\pi f_2t\right)=\frac{a}{2}\cos \left(\left(2\pi f_1-2\pi f_2\right)t\right)+\frac{a}{2}\cos \left(\left(2\pi f_1+2\pi f_2\right)t\right)& \mathrm{Eq}.\left(3\right)\\ -a\times \sin \left(2\pi f_1t\right)\sin \left(2\pi f_2t\right)=\frac{-a}{2}\cos \left(\left(2\pi f_1-2\pi f_2\right)t\right)+\frac{a}{2}\cos \left(\left(2\pi f_1+2\pi f_2\right)t\right)& \mathrm{Eq}.\left(4\right)\end{array}$

[0021] As shown through Equation (3) and Equation (4), the lower sideband (f₁−f₂) signal would be completely eliminated and only the upper sideband (f₁+f₂) signal remains. Therefore a single sideband signal results. In the same way, the upper sideband (f₁+f₂) signal could be eliminated and the lower sideband (f₁−f₂) signal remains such that a single sideband signal is generated through inverting one or three of the in-phase carrier S_Ir, the quadrature-phase carrier S_Qr, the in-phase component I_t, or the quadrature-phase component Q_t. Under this ideal condition, there is no IQ imbalance.

[0022] However, when the phase difference between the in-phase carrier S_Ir and the quadrature-phase carrier S_Qr is not 90°, or when their amplitudes are different, the lower sideband signal will not be eliminated as shown in Equation (3) and Equation (4) to result in an image signal. As mentioned above, if the quadrature-phase carrier S_Qr is given an additional phase shift θ, the mixed signal can then be represented as:

$\begin{array}{cc}-a\times \sin \left(2\pi f_1t\right)\sin \left(2\pi f_2t+\theta \right)=\frac{-a}{2}\cos \left(\left(2\pi f_1-2\pi f_2\right)t-\theta \right)+\frac{a}{2}\cos \left(\left(2\pi f_1+2\pi f_2\right)t+\theta \right)& \mathrm{Eq}.\left(5\right)\end{array}$

[0023] Therefore, in addition to the integrated upper sideband (f₁+f₂) signal in the transmitting signal S_t being shifted due to the additional phase shift θ, the integrated lower sideband (f₁−f₂) signal in the transmitting signal S_t is not eliminated, and the above-mentioned image signal results. Please refer to FIG. 3 and FIG. 4. FIG. 3 shows a diagram illustrating the relationship between the amplitude and frequency in an ideally mixed signal. FIG. 4 is a diagram illustrating the relationship between the amplitude and frequency in a mixed signal corresponding to a quadrature-phase carrier with an additional phase shift. As shown in FIG. 3, only the upper sideband (f₁+f₂) signal occurs in an ideally mixed signal. However, as shown in FIG. 4, an image signal occurs on the lower sideband (f₁−f₂) spectrum of the signal because of the additional phase shift.

[0024] Under non-ideal mixing as mentioned above, if the phase mismatch and the gain mismatch between these carriers are estimated correctly, distortion in demodulation caused by IQ imbalance can be compensated in a digital domain or in an analog domain. Since the calibration procedure for IQ imbalance is operated through the QAM transceiver 200 as shown in FIG. 2, the calibration unit 205 generates an input signal, where its in-phase component I_t is fed into the mixer 210 for up-conversion with the in-phase transmitting carrier S_It. Similarly, the quadrature-phase component Q_t of the input signal is fed into the mixer 220 for up-conversion with the quadrature-phase transmitting carrier S_Qt. The transmitting signal S_t is now transmitted to the loopback component 270 through the path selector 250. In this embodiment, the loopback component 270 provides an amplitude adjustment and a phase adjustment to adjust the transmitting signal S_t. This generates a loopback signal S_lb, which is then transmitted to the receiving end through the path selector 260. The output signal (i.e. the receiving signal S_r) of the path selector 260 is then transmitted to the mixers 230 and 240, where the in-phase receiving carrier S_Ir and the quadrature-phase receiving carrier S_Qr are used for down-converting the receiving signal S_r to generate the in-phase component I_r and the quadrature-phase component Q_r, respectively, which correspond to the output signal. Finally, the output signal is fed into the calibration unit 205 for performing a calibration operation to estimate calibration parameters for IQ imbalance. Please note that the loopback component 270 provides a plurality of amplitude adjustments and phase adjustments in sets, and selects one set from these sets to adjust the amplitude and the phase of the transmitting signal S_t.

[0025] In this section a commonly used IQ imbalance model is used for illustrating the estimation of the calibration parameters in the present invention. We assume that the in-phase transmitting carrier S_It, the quadrature-phase transmitting carrier S_Qt, the in-phase receiving carrier S_Ir, and the quadrature-phase receiving carrier S_Qr are represented as follows:

S_It=(1+α_t)cos(2πf₁t+θ_t)   Eq. (6)

S_Qt=−sin(2πf₁t)   Eq. (7)

S_Ir=(1+α_r)cos(2πf₁t+θ_r)   Eq. (8)

S_Qr=−sin(2πf₁t)   Eq. (9)

[0026] In Equations (6), (7), (8), and (9), the values α_t and α_r represent the amplitude error of the transmitting carrier and the amplitude error of the receiving carrier respectively, and the values θ_t and θ_r represent the phase errors of the transmitting carrier and the phase error of the receiving carrier respectively. The frequency f₁ is a carrier frequency, while the value t is a time index.

[0027] As shown in Equations (6), (7), (8), and (9), there are four error parameters that must be estimated. In this embodiment, since the loopback component 270 provides two sets of amplitude adjustment A and phase adjustment θ_d, there are eight unknown values to be solved for the input signal and the output signal. For a specific input signal, the calibration loop of the QAM transceiver 200 is capable of providing four equations. Thus, if the calibration unit 205 is able to provide two sets of different input signals or more, at least eight equations are obtained to successfully solve these eight unknown values. We assume that the in-phase transmitting carrier I_t and the quadrature-phase transmitting carrier Q_t are represented below as follows:

I_t=a×S   Eq. (10)

Q_t=b×S   Eq. (11)

[0028] where a and b are scalars of the amplitude, and S is any base-band or middle-band signal. Therefore, the transmitting signal S_t can be represented as follows:

S_t=(1+α_t)aS cos(2πf₁t+θ_t)−bS sin(2πf₁t)   Eq. (12)

[0029] The loopback signal S_lb outputted by the transmitting signal S_t through the loopback component 270 is then shown in the following:

S_lb=A[(1+α_t)aS cos(2πf₁t+θ_t−θ_d)−bS sin(2πf₁t−θ_d)]  Eq. (13)

[0030] The loopback signal S_lb passes through mixers 230, 240, with the low-frequency component of the loopback signal S_lb being extracted with a low pass filter (LPF, not shown in FIG. 2). The in-phase component I_r and the quadrature-phase component Q_r of the output signal can then be shown as follows:

I_r=LPF{A[(1+α_t)aS cos(2πf₁t+θ_t−θ_d)−bS sin(2πf₁t−θ_d)]×[(1+α_r)cos(2πf₁t+θ_r)]}=(½)A(1+α_t)(1+α_r)aS cos(θ_t−θ_d−θ_r)+(½)A(1+α_r)bS sin(θ_d+θ_r)   Eq. (14)

Q_r=LPF{A[(1+α_t)aS cos(2πf₁t+θ_t−θ_d)−bS sin(2πf₁t−θ_d)]×[−sin(2πf₁t)]}=(½)A(1+α_t)aS sin(θ_t−θ_d)+(½)AbS cos(θ_d)   Eq. (15)

[0031] Assumptions are made for parameters α_t, α_r, θ_t, and θ_r<<1 to simplify Equations (14) and (15) as shown below:

I_r=(½)AaS[(1+α_t+α_r)cos θ_d+sin θ_d(θ_t−θ_r)]+(½)AbS[(1+α_r)sin θ_d+cos(θ_dθ_r)]Eq.   (16)

Q_r=(½)AaS[−(1+α_t)sin θ_d +cos(θ_dθ_t)]+(½)AbS cos(θ_d)   Eq. (17)

[0032] Set values are substituted for (a, b) and (A, θ_d) respectively, with the resulting independent linear equations for each test set listed below:

(a, b)=(1, 0) and (A₁, θ_d1)

2I_r1/S=A₁[(1+α_t+α_r)cos θ_d1+sin θ_d](θ_t−θ_r)]  Eq. (18)

2Q_r1/S=A₁[−(1+α_t)sin θ_d1+cos(θ_d1θ_r)]  Eq. (19)

(a, b)=(0, 1) and (A₁, θ_d1)

2I_r2/S=A₁[(1+α_r)sin θ_d1+cos(θ_d1θ_r)]  Eq. (20)

2Q_r2/S=A₁ cos θ_d1   Eq. (21)

(a, b)=(1, 0) and (A₂, θ_d2)

2I_r3/S=A₂[(1+α_t+α_r)cos θ_d2+sin θ_d2(θ_t−θ_r)]  Eq. (22)

2Q_r3/S =A₂[−(1+α_t)sin θ_d2+cos(θ_d2θ_r)]  Eq. (23)

(a, b)=(0, 1) and (A₂, θ_d2)

2I_r4/S=A₂[(1+α_r)sin θ_d2+cos(θ_d2θ_r)]  Eq. (24)

2Q_r4/S=A₂ cos θ_d2   Eq. (25)

[0033] According to these above equations (18)-(25), all four error parameters including α_t, α_r, θ_t, and θ_r, are correctly calculated. These four error parameters can then be used to compensate for IQ imbalance using a prior art compensation scheme. This compensation scheme being well known to one skilled in the art is not discussed here for brevity. Please note that while the two sets of phase adjustments provided by the loopback component 270 are different, the above equations remain solved nonetheless. Moreover, when the difference between the two phase adjustments are 90°, the computation involved in solving these equations can be considerably simplified.

[0034] In practical applications, each set of amplitude and phase adjustments performed by the loopback component 270 corresponds to each individual path in the loopback component 270. That is, the amplitude adjustment and phase adjustment occurring for each path in the loopback component 270 varies. Therefore, different sets of amplitude and phase adjustments can be added into the transmitting signal S_t by selecting different paths. Please refer to FIG. 5. FIG. 5 is a diagram of an embodiment of the loopback component 270 shown in FIG. 2. As shown in FIG. 5, the loopback component 270 includes a plurality of paths and a selector 272, wherein the selector 272 is used to select one path from a plurality of paths. Each path corresponds to a specific amplitude adjustment A₁˜A_n and phase adjustments θ₁˜θ_n. In another embodiment of the loopback component 270, the selector 272 is placed in the output end of the loopback component 270. In this case, the transmitting signal S_t is fed into all paths, being adjusted by all the various amplitude adjustments A₁˜A_n and phase adjustments θ₁·θ_n, and a specific path is chosen through the selector 272 for passing a desired adjusted transmitting signal S_t to the receiving end of the path selector 260.

[0035] Please refer to FIG. 6. FIG. 6 is a diagram of an embodiment of the path selector 250 shown in FIG. 2. As shown in FIG. 6, the path selector 250 comprises a coupler implemented through two neighbor leads. When the transmitting signal S_t is fed into the path selector 250 a portion of energy is coupled and outputted to the coupling end. To ensure that the portion of coupled energy does not influence normal operation of the receiving end in the QAM transceiver 200, a switch (not shown) can be additionally set in the input end of the loopback component 270. The switch can be switched off while the QAM transceiver 200 performs normal operation, and turned on while the QAM transceiver 200 performs the calibration operation. Please note that the use of the switch largely depends on practical design requirements. Some implementations of QAM transceivers are such that the receiving end and the transmitting end do not operate simultaneously, where only one end operates at a time. In this situation, the switch is no longer necessary in the operation of the QAM transceiver 200. Please note that the use of the switch is just for an example and shall not be a limitation to the present invention. In another embodiment, the path selector 250 is implemented using a simple selector. The selector can couple the transmitting signal S_t to the antenna 280 while the QAM transceiver 200 undergoes the normal operation. The transmitting signal S_t can also be coupled to the loopback component 270 by the selector while the QAM transceiver 200 undergoes the calibration operation.

[0036] The QAM transceiver and the method for calibrating IQ imbalance disclosed in the present invention utilize a loopback component to provide two or more sets of amplitude and phase adjustments such that the outputted transmitting signal is accurately adjusted. The adjusted transmitting signal is then received and fed into the receiving end to estimate the error parameters for IQ imbalance. According to the present invention, additional circuits used in the present invention, i.e. the loopback component, reduce the occurrence of errors found in the prior art, and the cost function is thus not required for evaluating IQ imbalance, so stability will no longer be an issue to be considered.

[0037] Those skilled in the art will readily observe that numerous modifications and alterations of the device and method may be made while retaining the teachings of the invention. Accordingly, the above disclosure should be construed as limited only by the metes and bounds of the appended claims.