Termination resistance independent system for impedance matching in high speed input-output chip interfacing

[0001] The present application is a continuation-in-part of U.S. patent application Ser. No. 09/255,997, filed on Feb. 23, 1999, which is pending and is assigned to the assignee of the subject application.

[0002] The present invention relates generally to fabrication of electronic devices and, more particularly, to a system and structure for impedance matching in very high speed input-output (I/O) chip interfacing to improve performance and reliability of I/O chips and systems that contain I/O chips.

[0003] One of the most important aspects in integrated chip system design is the interface considerations for very short channel Complementary Metal-Oxide-Semiconductor (CMOS) devices in high-speed applications. Beginning with the driving circuit, an on-chip voltage swing of ground to Vdd (internal power supply) must be communicated off-chip to external devices. The path from the driver output to the output pad involves capacitive coupling effects to other on-chip signals. Moreover, the chip packaging system adds inductances to the circuit. The external signals must then traverse some interconnect (or transmission line), whether card traces or multi-chip module (MCM) connections. The signals are subject to additional deformations at this point due to transmission line effects introduced by the interconnects. The use of external decoupling capacitors to stabilize the card power supply provided to the chip is known in the art. For low-impedance card power plane connections, these external capacitors do not reduce the on-chip simultaneous switching noise of the driver circuit.

[0004] Transmission line effects become significant when the round trip propagation delay from the sending chip to the receiving chip is greater than the rise time of the transmitted signal. This condition is almost always met for modern CMOS-based digital systems. When this is the case, reflections occur on the signal line due to impedance mis-matches between the source, the transmission line, and the load. These reflections are superimposed on the transmitted signal, causing significant overshoot, undershoot, and system-wide noise. These conditions can cause both performance problems and reliability problems.

[0005] The performance problem is a reduction in the valid-data window of transmitted pulses with respect to the system clock. The reliability problem involves devices with ultra-thin gate dielectric layers, less than 5 nm thickness, for which overshoots and undershoots can have a significant effect in reducing the effective useful life of the thin gate dielectric. This can lead to catastrophic breakdown. The voltage overshoots and undershoots can also cause latch-up in CMOS devices leading to serious reliability problems. For high speed CMOS applications where the data rates are in the range of 1 GHz to 10 GHz and above, these performance and reliability problems can cause severe design limitations and prove costly.

[0006] The deficiencies of conventional, high-speed input-output interfaces show that a need exists for improvement. To overcome the shortcomings of conventional systems, a new system and structure for impedance matching in high-speed input-output chip interfacing is provided. It is an object of the present invention to provide a system and structure to achieve impedance matching at a driver circuit output, thereby preventing and not merely compensating for high voltage transients (overshoots) and low voltage transients (undershoots). It is another object of the present invention to provide a system and structure which can be manually adjusted to provide impedance matching at a driver circuit output during system set-up. It is yet another object of the present invention to provide an economical and manufacturable system and structure which can be manually adjusted to provide impedance matching at a driver circuit output for actual operating conditions of a specific system.

[0007] It is still another object of the present invention to provide a system and structure capable of automatically adjusting to provide impedance matching at a driver circuit output when operating conditions cause changes to the system. Another object of the present invention is to provide a control circuit for an impedance matching system which detects overshoots and undershoots on the driver circuit output and automatically provides a control voltage to an adjustment mechanism to adjust the length of an adjustable-length transmission line providing impedance matching based on actual overshoots and undershoots. Still another object of the present invention is to provide an impedance matching system capable of performing its functions under a wide range of values for the various components comprising both the on-chip driver circuit and the off-chip interface circuitry.

[0008] To achieve these and other objects, and in view of its purposes, the present invention provides a process and structure for an impedance matching network. The structure includes an adjustable-length transmission line located between the driver circuit and the receiver circuit (load) on the system printed circuit board (card). The adjustable-length transmission line is comprised of two conductive stripes connected in parallel with an input-output driver circuit at one end, electrically isolated at the opposite end, and short circuited by a movable stub intermediate of the ends. The adjustable length transmission line is located on the system printed circuit board between the driver circuit and the receiver circuit. Optionally, the impedance matching network may further comprise an adjustment mechanism and a control circuit.

[0009] The impedances at the driver circuit output are matched by adjusting the length of the adjustable-length transmission line such that the composite reactance of the load, the transmission line between the driver circuit chip and the load, the connections, and the adjustable-length transmission line equals the converse of the driver reactance. In a first embodiment, the length of the adjustable-length transmission line can be set or adjusted manually at system set-up, such as by a systems engineer or operator. In order to adjust the adjustable-length transmission line, the systems engineer or operator observes the waveform using electronic measuring equipment such as an oscilloscope. The position of the moveable stub is manually adjusted until the magnitude of undershoots and overshoots as viewed on the electronic measuring equipment is minimized.

[0010] In a second embodiment, the length of the adjustable-length transmission line is adjusted by the adjustment mechanism. The adjustment mechanism comprises an elecromagnetic coil, a moveable magnetic core, and a return spring. The electromagnetic coil creates a magnetic field proportional to a control current from the control circuit. The magnetic field provides a force to the moveable magnetic core acting in a first direction. The return spring provides a force to the moveable magnetic core acting in a second direction opposite to the first direction. The moveable magnetic core is attached to the moveable stub whose position determines the length of the adjustable-length transmission line.

[0011] The control circuit generates a control current which varies in response to high and low transient voltages on the driver circuit output. In the second embodiment, the control circuit comprises a positive detect circuitry and a negative detect circuitry, each of which provides a high (up) pulse to a counter when triggered by a transient, and a CLK down generator which provides a low (down) pulse to the counter as a function of the system clock. Both the positive detect circuitry and the negative detect circuitry are self-resetting.

[0012] In the second embodiment, the control circuit further comprises a quantity of n N-type field effect transistors connected in parallel between the adjustment mechanism of the adjustable length transmission line and the system ground. The transistors have control gates connected to an n-bit counter which is incrementally increased by the positive detect circuitry when an overshoot occurs on the driver circuit output and by the negative detect circuitry when an undershoot occurs on the driver circuit output. Thus, the number of transistors drawing current and the resulting control current are adjusted until no transients are occurring on the driver circuit output.

[0013] The present invention provides considerable improvement over the prior art. Whereas the prior art has been directed toward compensating for transient voltages using approaches such as isolation rings, forward biasing, excess dielectric, and decoupling capacitors on the system card, the present invention is directed to preventing voltage transients on the driver circuit output by impedance matching. The present invention can be economically fabricated and implemented using current process technologies and existing input-output semiconductor devices. In addition, by adjusting the length of the adjustable-length transmission line based on the magnitude of the transients in a particular system, the present invention can operate in systems having a wide range of operating conditions and in systems having varying operating conditions. For instance, when operating conditions change dynamically during operation, the present invention can automatically adjust the impedance to provide matching. The present invention achieves these benefits in the context of known process technology.

[0014] It is to be understood that both the foregoing general description and the following detailed description are exemplary, but are not restrictive, of the invention.

[0015] The features and advantages of the impedance matching network according to the present invention and further details of the process and structure for such an impedance matching network are best understood from the following detailed description when read in connection with the accompanying drawing. It is emphasized that, according to common practice, the various features of the drawing are not to scale. On the contrary, the dimensions of the various features are arbitrarily expanded or reduced for clarity. Included in the drawing are the following figures:

[0016] FIG. 1 illustrates a conventional circuit model for a driver circuit, an off-chip receiver circuit, and a transmission line;

[0017] FIG. 2A illustrates a circuit model for the driver circuit, the impedance matching network of the present invention, the transmission line, the off-chip receiver circuit, and connectionsshowing the impedance matching network as a black box;

[0018] FIG. 2B illustrates the conductive lines of the transmission line;

[0019] FIG. 3A illustrates the adjustable-length transmission line of the matching network according to the first embodiment of the present invention;

[0020] FIG. 3B illustrates the adjustable-length transmission line of the matching network and the adjustment mechanism of the matching network (shown in the relaxed state in which the length of the adjustable-length transmission line is at its maximum) according to a second embodiment of the present invention, in which the magnetic coil is shown cut open to highlight the magnetic core;

[0021] FIG. 4 illustrates an alternate circuit model showing the impedance matching network as an adjustable length transmission line, according to the present invention;

[0022] FIG. 5 illustrates a circuit model showing a composite impedance as derived for the receiver side of the matching network;

[0023] FIG. 6A shows a block diagram of the control circuit according to the second embodiment of the present invention;

[0024] FIG. 6B shows a schematic of the positive detect circuit according to the second embodiment of the present invention;

[0025] FIG. 6C shows a schematic of the negative detect circuit according to the second embodiment of the present invention;

[0026] FIG. 7 shows the calculat e d result for impedance matching for the cases in which selected values of load resistance (termination resistance) range from 50 ohms to 500 mega ohms, with the results provided in terms of the length for the adjustable length transmission line required to achieve impedance matching, as a function of frequency ranging from 0.5 to 100 giga hertz (GHz);

[0027] FIG. 8 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the receiver inductance ranging from 1 to 8 nano henry (nH);

[0028] FIG. 9 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the driver inductance ranging from 1 to 16 nH;

[0029] FIG. 10 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the receiver capacitance ranging from 1 pF to 12 pF;

[0030] FIG. 11 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the driver-capacitance ranging from 1 pF to 12 pF;

[0031] FIG. 12 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the dielectric constant for both the transmission line and the adjustable-length transmission line ranging from 2 to 6;

[0032] FIG. 13 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the length of the transmission line ranging from 51 to 127 mm (2 to 5 inches); and

[0033] FIG. 14 shows the length of the adjustable-length transmission line required for impedance matching, as a function of frequency, for selected values of the characteristic impedance of both the transmission line and the adjustable-length transmission line in the range from 20 ohms to 200 ohms.

[0034] The present invention will now be described in detail with reference to the accompanying drawing, in which like reference numbers designate similar or corresponding elements, regions, and portions. Several processes of compensating for the overshoots and undershoots caused by impedance mis-match are known in the art. Guard rings, excess dielectric, and forward biasing are some examples of the techniques used to compensate for the transient voltages caused by impedance mis-match. The present invention provides a process for matching the reactive component of the impedance of the load and the transmission line to the impedance of the driver circuit, thereby preventing overshoots and undershoots and their associated performance and reliability problems.

[0035] FIG. 1 shows a circuit model for a driver circuit (20), an off-chip receiver circuit (10) and a transmission line (40) according to the prior art. When the net reactance at the output of the driver circuit (20) is not zero (impedance mis-match), reflections occur on the signal line. These reflections cause overshoots and undershoots.

[0036] FIG. 2A shows a circuit model of the most significant resistive and reactive (e.g., inductive and capacitive) components involved in a system where a driver circuit (20) on one chip is used to supply a signal (voltage) to a receiver circuit (10) on a different chip from the driver circuit. The signal is transmitted from the driver circuit (20) to the receiver circuit (10) on the transmission line (40). An impedance matching network (50), which is the focus of the present invention, is located between the driver circuit (20) and the transmission line (40).

[0037] The on-chip driver circuit (20) has a driver resistance (R20) and a driver capacitance (C20). The driver resistance (R20) is typically about 50 ohms. The driver capacitance (C20) is typically about 4.7 pF. As will be shown later, however, the present invention can function with values that differ from the given typical values.

[0038] The receiver circuit (10), which is typically not but might be on the same chip as the driver circuit (20), has a receiver resistance (R10) and a receiver capacitance (C10). The receiver resistance (R10) can vary in value depending on the particular system with a typical value being about 50 ohms. It will be demonstrated that the present invention will work independently of the termination resistance of the receiver circuit (10) from a receiver resistance (R10) of 50 ohms to a receiver resistance (R10) approaching the infinite resistance of an open circuit. The receiver capacitance (C10) is typically about 3.7 pF.

[0039] The connections at the receiver circuit (10) provide a receiver inductance (L31). Chip packaging connections at the driver circuit (20) provide a driver inductance (L32). The receiver inductance (L31) and the driver inductance (L32) are typically each about 2 nH.

[0040] As shown in FIG. 2B, the transmission line (40) is composed of printed, conductive stripes (42) of conductive material on a system circuit card (30) with a length (D40) of about 76 mm (3 inches). The stripes (42) are separated by an insulator (44). The dielectric constant (40) of the insulator (44) between the stripes (42) is typically about 4.5. The relative permeability (40) of the insulator (44) is typically about 1. The characteristic impedance (Z40) of the transmission line (40) is typically about 50 ohms.

[0041] Unless otherwise specified, all calculations are performed and all graphs are made using the typical values provided above for driver resistance (R20), driver capacitance (C20), receiver resistance (R10), receiver capacitance (C10), driver inductance (L32), receiver inductance (L31), transmission line length (D40), transmission line dielectric constant (40), transmission line relative permeability (40), and transmission line characteristic impedance (Z40). As will be demonstrated later, however, the present invention can provide impedance matching over a wide range for each of these characteristics.

[0042] First Embodiment of the Present Invention

[0043] In a first embodiment, the matching network (50) comprises an adjustable-length transmission line (52) having a variable length (D52). The adjustable-length transmission line (52), as shown in FIG. 3A, comprises two parallel conductive lines (54) (e.g., stripes or circuit traces) located on the system card (30), connected in parallel with an on-chip driver circuit (20) at one end, electrically isolated at the opposite end, and shorted by a movable stub (56) intermediate of the ends of the parallel conductive lines (54). The conductive lines (54) can be composed of a variety of conductive materials, such as aluminum, gold, or most preferably copper. The conductive lines (54) preferably have a width of between about 0.5 inches and one inch, a thickness of between about 400 m and 800 m, and are preferably spaced apart a distance of between about 0.5 inches and 2 inches.

[0044] The movable stub (56) can be composed of a variety of conductive materials, including aluminum, copper, or most preferably stainless steel. The movable stub (56) is preferably formed using conventional micro-machining techniques. The movable stub (56) simultaneously contacts both of the parallel conductive lines (54), and has a thickness of between about 3 mm and 10 mm. The movable stub (56) contacts the parallel conductive lines (54) with a pressure of between about 5 grams and 20 grams.

[0045] The movable stub (56) further comprises a mechanism to adjust the position along the length of the parallel conductive lines (54) at which the movable stub (56) contacts the parallel conductive lines (54), thereby changing the length (D52) of the adjustable-length transmission line (52). The position of the movable stub (56) can be adjusted in a number of ways, including but not limited to direct manual manipulation of the movable stub (56) and indirect manipulation of the movable stub (56) through a vernier device (58) as is known in the art. A vernier device is a short, graduated scale that slides along a longer, graduated instrument and is used to indicate fractional parts of divisions, as in a micrometer.

[0046] In the preferred process, the movable stub (56) is initially positioned along the length of the parallel conductive lines (54) such that the adjustable-length transmission line (52) is set to its maximum length (shorted as far as possible from the connection of the parallel conductive lines (54) to the output of the driver circuit (20)). The output of the driver circuit (20) is monitored using measuring equipment such as an oscilloscope to determine the magnitude of overshoots and undershoots. If the magnitude of overshoots and undershoots exceeds an acceptable threshold for the system, the position of the movable stub (56) is adjusted incrementally to increase the length (D52) of the adjustable-length transmission line (52). Most preferably, the output of the driver circuit (20) is monitored while the length (D52) of the adjustable-length transmission line (52) is simultaneously adjusted until the magnitude of overshoots and undershoots is within the acceptable threshold. Care must be taken in choosing the monitoring equipment to minimize the impedance-loading effect introduced by the monitoring equipment. The acceptable threshold for overshoot and undershoot magnitude depends upon the particular system based on performance, system set-up, and reliability requirements for the particular system.

[0047] Second Embodiment of the Present Invention

[0048] In a second embodiment, the matching network (50) comprises an adjustable-length transmission line (52) having a variable length (D52), an adjustment mechanism (60), and a control circuit (70). For ease of manufacturing and economic implementation, the matching network is preferably located on the system card (30). This allows for use of existing I/O chips and manufacturing methods for conductive lines and a moveable stub on a system card which are known in the art.

[0049] As shown in FIG. 3B, the adjustable-length transmission line (52) comprises two parallel conductive lines (54) located on the system card (30), connected in parallel with an on-chip driver circuit (20) at one end, electrically isolated at the opposite end, and shorted by a movable stub (56) intermediate of the two ends of the parallel conductive lines (54). The conductive lines (54) can be composed of a variety of conductive materials, such as aluminum, gold, or most preferably copper. The conductive lines (54) preferably have a width of between about 0.1 and 0.25 inches, a thickness of between about 400 and 800 m, and are preferably spaced apart a distance of between about 0.1 and 0.25 inches.

[0050] The movable stub (56) can be composed of a variety of conductive materials, including aluminum, copper, or most preferably stainless steel. The movable stub (56) is preferably formed using conventional micro-machining techniques. The movable stub (56) simultaneously contacts both of the parallel conductive lines (54) and has a thickness of between about 3 and 10 mm. The movable stub (56) contacts the parallel conductive lines (54) with a pressure of between about 20 and 40 grams/cm². The moveable stub (56) is attached to a magnetic core (68) which is part of the adjustment mechanism (60).

[0051] The adjustment mechanism (60) receives a control current from the control current node (71) of the control circuit (70) causing it to transfer a linear motion to the moveable stub (56). The linear movement of the moveable stub (56) adjusts the length (D52) of the adjustable-length transmission line (52). The adjustment mechanism (60) is located adjacent to the adjustable-length transmission line (52) on the system card (30).

[0052] As shown in FIG. 3B, the adjustment mechanism (60) preferably comprises an electromagnetic coil (62) having between 50 and 300 turns and preferably 100 turns of gold, aluminum, or most preferably copper wire. The copper wire preferably has a diameter of between about 0.2 and 0.5 mm. The electromagnetic coil (62) is connected at one end to a power supply (64) and at the other end to the control current node (71) of the control circuit (70). The electromagnetic coil (62) creates a magnetic field that provides a force acting in a first direction on the magnetic core (68) located inside of the electromagnetic coil (62). The magnetic core (68) is attached such that it has only one degree of freedom parallel to the axis of the electromagnetic coil (62). Most preferably, the magnetic force created by the electromagnetic coil (62) acting on the magnetic core (68) causes the length (D52) of the adjustable-length transmission line (52) to decrease. A return spring (66) fixedly attached to the system card (30) on one end and attached to the magnetic core (68) on the opposite end provides a force on the magnetic core (68) acting in a second direction opposite to the first direction.

[0053] When the number of energized NFETs (80) increases due to transients on the output of the driver circuit (20), the control current through the electromagnetic coil (62) increases. As the control current through the electromagnetic coil (62) increases, the electromagnetic force acting on the magnetic core (68) increases, causing the magnetic core (68) to move in the first direction, decreasing the length (D52) of the adjustable-length transmission line (52).

[0054] When the number of energized NFETs (80) decreases due to the clock down generator (83), the control current through the electromagnetic coil (62) decreases. As the control current through the electromagnetic coil (62) decreases, the return spring (66), which is in tension, causes the magnetic core (68) to move in the second direction, increasing the length (D52) of the adjustable-length transmission line (52).

[0055] Referring to FIG. 6A, the control circuit (70) detects overshoots or (positive) high voltage transients and undershoots or (negative) low voltage transients on the output of the driver circuit (20) and provides a control current in the control current node (71) to the adjustment mechanism (60). The control current in control current node (71) causes the adjustment mechanism (60) to adjust the length (D52) of the adjustable-length transmission line (52), thereby minimizing overshoots and undershoots.

[0056] The control circuit (70), as shown in FIG. 6A, comprises positive detect circuitry (72) and negative detect circuitry (74) connected to the output of the driver circuit (20). When an overshoot occurs on the output of the driver circuit (20), the positive detect circuitry (72) detects the overshoot and generates a high pulse at node UP1 (77). When an undershoot occurs on the output of the driver circuit (20), the negative detect circuitry (74) detects the undershoot and generates a high pulse at node UP2 (79).

[0057] Nodes UP1 (77) and UP2 (79) are connected to the inputs of an OR gate (76), which is in turn connected at its output to an n-bit counter (78). The n-bit counter (78) is connected at its outputs to the control gates of a quantity n N-type field effect transistors or NFETs (80) which are connected in parallel. The quantity n is large enough to provide adjustment increments that will meet the overshoot-undershoot threshold established for the system design, preferably between about 4 and 16. A quantity of these NFETs (80) are made active (energized or turned on) equal to the count of the n-bit counter (78). A current is drawn through the NFETs (80) which is proportional to the number of NFETs (80) which are energized. The NFETs (80) are connected to ground at their sources and to the node CONTROL CURRENT (71) at their drains. The node CONTROL CURRENT (71) is connected to the adjustment mechanism.

[0058] The control current at the node CONTROL CURRENT (71) is determined by the total currents in the n NFETs which are energized. The node CONTROL CURRENT (71) is connected to the adjustment mechanism (60), providing the control current to the adjustment mechanism (60).

[0059] A down signal is periodically generated by the CLK down generator (83) on the node DOWN (85) as a function of the system clock (88). The down signal causes the n-bit counter (78) to count down, thereby reducing by one the quantity of NFETs (80) which are energized. The periodic down signal prevents the impedance matching network from being over-adjusted because of impedance changes due to dynamic environmental conditions such as temperature.

[0060] When a positive transient or overshoot occurs on the output of the driver circuit (20), the positive detect circuitry (72) generates a high pulse at node UP1 (77). The positive detect circuitry (72) is preferably implemented as shown in FIG. 6B. A pulse up transistor (T100), which is an N-well PFET, is connected to the driver circuit power supply (101) at its body and to a high reference potential, typically the driver circuit power supply (101), at its control gate (VGATE). The pulse up transistor (T100) is connected to the output of the driver circuit (20) at its source and to node PULSE (103) at its drain. When a positive transient occurs on the output of the driver circuit (20), the pulse up transistor (T100) switches the node PULSE (103) from low to high. An up interrogate transistor (T104), which is an NFET, is connected to the node PULSE (103) at its control gate, to ground at its source, and to node UPC1 (105) at its drain. When node PULSE (103) switches to high, the up interrogate transistor (T104) switches node UPC1 (105) from high to low.

[0061] Nodes UPC1 (105) and DOWN (85) are connected to the inputs of a NOR gate (G106) which generates an up (high) pulse on node UP1 (77) when node UPC1 (105) is low due to a high transient, provided node DOWN (85) has not been switched to a high level by the CLK down generator (83). Node UPC1 (105) is also connected to a voltage keeper comprising an inverter (I108) connected to the control gate of a PFET keeper transistor (T110). The source of the keeper transistor (T110) is connected to internal power and the drain is connected to node UPC1 (105) such that node UPC1 (105) is held at a high state until interrogation by the up interrogate transistor (T104) can occur, thereby preventing accidental tripping by system noise.

[0062] Node UPC1 (105) is also connected to a reset loop comprising an odd number of reset delay inverters (I112) connected in series. The last reset delay inverter (I112) is connected to node RESET (109). The number of reset delay inverters (I112) is sufficient to cause a delay longer than the duration of the transient. Experimental modeling has shown that seven reset delay inverters (I112) are typically sufficient. When node UPC1 (105) switches to low, the reset loop switches node RESET (109) from low to high. The node RESET (109) is connected to the control gate of a pulse reset transistor (T114), which is an NFET connected to ground at its source and to node PULSE (103) at its drain. When node RESET (109) switches to high, the pulse reset transistor (T114) switches node PULSE (103) back to its original state of low.

[0063] Node RESET (109) is also connected to a UPC1 reset inverter (I116), which is in turn connected to the control gate of a UPC1 reset transistor (T118). The UPCl reset transistor (T118) is a PFET connected to internal power at its source and to node UPC1 (105) at its drain. When node RESET (109) switches to high, the UPC1 reset transistor (T118) switches node UPC1 (105) back to its original state of high.

[0064] A first optional isolating resistive device (R120) can be connected between internal power and node RESET (109) to prevent accidental switching due to coupling to external signals or system noise. Likewise, a second optional isolating resistive device (R122) can be connected between ground and node PULSE (103). The isolating resistive devices (R120, R122) can be resistors or resistively connected FETs.

[0065] The negative detect circuitry (74) is a mirror image of the positive detect circuitry (72). When a negative transient or undershoot occurs on the output of the driver circuit (20), the negative detect circuitry (72) generates a high pulse at node UP2 (79). The negative detect circuitry (72) is preferably implemented as shown in FIG. 6C.

[0066] A pulse down transistor (T200), which is an NFET, is connected to a low reference potential, typically ground, at its control gate (VGATE2). The pulse down transistor (T200) is connected to the output of the driver circuit (20) at its source and to node PULSE DOWN (203) at its drain. When a negative transient occurs on the output of the driver circuit (20), the pulse down transistor (T200) switches the node PULSE DOWN (203) from high to low. A down interrogate transistor (T204), which is a PFET, is connected to the node PULSE DOWN (203) at its control gate, to internal power at its source, and to node UPC2 (205) at its drain. When the node PULSE DOWN (203) switches to low, the down interrogate transistor (T204) switches node UPC2 (205) from low to high.

[0067] Down (85) is connected to the input of an inverter (I240) whose output is connected to an input of a NAND gate (G206). Node UPC2 (205) is connected to the other input of NAND gate (G206). NAND gate (G206) is connected in series with a UP2 inverter (I230) then a node UP2 (79). An up (high) pulse is generated on node UP2 (79) when node UPC2 (205) is high due to a low transient, provided node DOWN (85) has not been switched to a high level by the CLK down generator (83). Node UPC2 (205) is also connected to a voltage keeper comprising an inverter (I208) connected to the control gate of an NFET keeper transistor (T210). The source of the NFET keeper transistor (T210) is connected to ground and the drain is connected to node UPC2 (205) such that node UPC2 (205) is held at a low state until interrogation by the down interrogate transistor (T204) can occur, thereby preventing accidental tripping by system noise.

[0068] Node UPC2 (205) is also connected to a second reset loop comprising an odd number of reset delay inverters (I212) connected in series. The last reset delay inverter (I212) is connected to node RESET2 (209). The number of reset delay inverters (I212) is sufficient to cause a delay longer than the duration of the transient. Experimental modeling has shown that seven reset delay inverters (I212) are typically sufficient. When node UPC2 (205) switches to high, the reset loop switches node RESET2 (209) from high to low. The node RESET2 (209) is connected to the control gate of a pulse down reset transistor (T214), which is a PFET connected to internal power at its source and to node PULSE DOWN (203) at its drain. When node RESET2 (209) switches to low, the pulse down reset transistor (T214) switches node PULSE DOWN (203) back to its original state of high.

[0069] Node RESET2 (209) is also connected to a UPC2 reset inverter (I216), which is in turn connected to the control gate of a UPC2 reset transistor (T218). The UPC2 reset transistor (T218) is an NFET connected to ground at its source and to node UPC2 (205) at its drain. When node RESET2 (209) switches to low, the UPC2 reset transistor (T218) switches node UPC2 (205) back to its original state of low.

[0070] A third optional isolating resistive device (R220) can be connected between ground and node RESET2 (209) to prevent accidental switching due to coupling to external signals or system noise. Likewise, a fourth optional isolating resistive device (R222) can be connected between internal power and node PULSE DOWN (203). The isolating resistive devices (R220, R222) can be resistors or resistively connected FETs.

[0071] In the above description, the terms low pulse, low state, and low and, similarly, the terms high pulse, high state, and high refer to relative voltage states and do not imply a specific magnitude.

[0072] Impedance Matching Calculations

[0073] The purpose of the adjustable-length transmission line (52) is to reduce transient voltages by providing a matching impedance for the reactive component of the impedance of the receiver circuit (10) to the driver circuit (20). When the adjustable-length transmission line (52) is short circuited at one end, the input impedance at the other end will have no resistive component. The reactive component (i.e., capacitive or inductive) of the input impedance will have a magnitude dependent on the length (D52) of the adjustable-length transmission line (52) and the frequency (f) of the signal from the driver circuit (20).

[0074] Thus, with a reactive impedance that can be adjusted, impedance matching between the on-chip driver circuit (20) and the off-chip receiver circuit (10) can be achieved for a wide range of configurations. Frequency (f) is dependant upon system design, where performance demands increasingly require operation at ultra-high frequencies (0.5 GHz and above). The required length (D52) of the adjustable-length transmission line (52) can be calculated using electromagnetic principles as described in chapter 12 of Engineering Electromagnetics, by William H. Hayt, Jr., McGraw-Hill, 1981, incorporated herein by reference.

[0075] Referring again to FIG. 2A, the resistive component (R1) and the reactive component (X1) of the input impedance (Z1) on the receiver side of the transmission line (40) are given by Equations 1 and 2, respectively, in terms of frequency (f), receiver resistance (R10), receiver capacitance (C10), and receiver inductance (L31). As is known in the art, impedance can generally be expressed as ZRjX and, therefore, the input impedance (Z1) can be expressed in the form of Equation 0.

[0076] Equations 1 and 2 can be derived for an inductor (such as receiver inductance (L31)) in series with a parallel capacitor (such as receiver circuit capacitance (C10)) and resistor (such as receiver circuit resistance (R10)), as shown in FIG. 2A. Angular frequency () is equal to 2f. $\begin{array}{cc}\mathrm{R1}=\frac{\mathrm{R10}}{1+{\left(\mathrm{R10}\mathrm{C10}\right)}^2}& \left(\mathrm{Equation}1\right)\\ \mathrm{X1}=\frac{\begin{array}{c}\mathrm{L31}-\left({\mathrm{R10}}^2\mathrm{C10}\right)+\\ \left({}^3{\mathrm{R10}}^2\mathrm{L31}{\mathrm{C10}}^2\right)\end{array}}{1+{\left(\mathrm{R10}\mathrm{C10}\right)}^2}& \left(\mathrm{Equation}2\right)\end{array}$

[0077] From Engineering Electromagnetics, the input impedance (Z4) at the driver side of the transmission line (40) can be expressed in terms of the characteristic impedance (Z40) of the transmission line (40), the load or input impedance (Z1), the transmission line length (D40), and a phase factor (40) for the transmission line (40). $\begin{array}{cc}\mathrm{Z4}=\mathrm{Z40}\frac{\mathrm{Z1}+\left[j\mathrm{Z40}\tan \left(40\mathrm{D40}\right)\right]}{\mathrm{Z40}+\left[j\mathrm{Z1}\tan \left(40\mathrm{D40}\right)\right]}& \left(\mathrm{Equation}3\right)\end{array}$

[0078] The phase factor (40) is dependant upon the frequency (f) of the signal from the driver circuit (20) and the dielectric constant (40) and the relative permeability (40) for the transmission line (40), and can be determined using equations 4A and 4B, where c is the speed of light (310⁸ m/s) $\begin{array}{cc}40=\frac{2f}{\mathrm{v40}}& \text{(Equation 4A)}\\ \mathrm{v40}=\frac{c}{{\left(4040\right)}^{-0.5}}& \text{(Equation 4B)}\end{array}$

[0079] The resistive component (R4) and the reactive component (X4) of the input impedance (Z4) on the driver side of the transmission line (40) can be determined by substituting Equation 0 into Equation 3 and separating the real (resistive) and imaginary (reactive) components. The resistive component (R4) of the input impedance (Z4) at the driver side of the transmission line (40) can be expressed as Equation 5, and the reactive component (X4) of the input impedance (Z4) at driver side of the transmission line (40) can be expressed as Equation 6. $\begin{array}{cc}\mathrm{R4}={\left(\mathrm{R1}\mathrm{Z40}\right)}^2\frac{1+{\tan }^2\left(40\mathrm{D40}\right)}{\begin{array}{c}{\left[\mathrm{Z40}-\left(\mathrm{X1}\tan \left(40\mathrm{D40}\right)\right)\right]}^2+\\ {\left[\mathrm{R1}\tan \left(40\mathrm{D40}\right)\right]}^2\end{array}}& \left(\mathrm{Equation}5\right)\\ \mathrm{X4}=\mathrm{Z40}\frac{\begin{array}{c}\{\left[\mathrm{X1}+\left(\mathrm{Z40}\tan \left(40\mathrm{D40}\right)\right)\right]\\ \left[\mathrm{Z40}-\left(\mathrm{X1}\tan \left(40\mathrm{D40}\right)\right)\right]\}-\\ \left[{\mathrm{R1}}^2\tan \left(40\mathrm{D40}\right)\right]\end{array}}{\begin{array}{c}{\left[\mathrm{Z40}-\left(\mathrm{X1}\tan \left(40\mathrm{D40}\right)\right)\right]}^2+\\ {\left[\mathrm{R1}\tan \left(40\mathrm{D40}\right)\right]}^2\end{array}}& \left(\mathrm{Equation}6\right)\end{array}$

[0080] As shown in FIG. 2A, R4 and X4 are also the resistive and reactive components of the impedance at the receiver side of the matching network (50). The driver circuit (20), matching network (50), transmission line (40) and receiver circuit (10) of FIG. 2A can now be reduced to the circuit shown in FIG. 4, where the impedance (Z50) of the matching network (50) and the input impedance (Z4) on the receiver side of the matching network (50) are in parallel.

[0081] The equivalent impedance (Z50) of the matching network (50) has no resistive component because the adjustable-length transmission line (52) is short circuited. The equivalent impedance (Z50) of the matching network (50), as given by Equation 7, can be expressed as a function of the characteristic impedance (Z52) of the adjustable-length transmission line (52), the phase factor (52) of the adjustable-length transmission line (52), and the length (D52) to which the adjustable-length transmission line (52) is set.

[0082] The phase factor (52) for the adjustable-length transmission line (52) is dependent upon the frequency (f) of the signal from the driver circuit (20) and the dielectric constant (52) and the relative permeability (52) for the adjustable-length transmission line (52). The phase factor (52) for the adjustable-length transmission line (52) can be determined using equations 8A and 8B, where c is the speed of light (310⁸ m/s). $\begin{array}{cc}52=\frac{2f}{\mathrm{v52}}& \text{(Equation 8A)}\\ \mathrm{v52}=\frac{c}{{\left(\mathrm{52}52\right)}^{-0.5}}& \text{(Equation 8B)}\end{array}$

[0083] Referring to FIG. 5, the net input impedance (Z3) at the driver side of the matching network (50) is given by Equation 9. The resistive component (R3) of the net input impedance (Z3) is given by Equation 10. The reactive component (X3) of the net input impedance (Z3) is given by Equation 11. $\begin{array}{cc}\mathrm{Z3}=\frac{\left[j\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]\left(\mathrm{R4}+j\mathrm{X4}\right)}{\mathrm{R4}+j\mathrm{X4}+\left[j\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]}& \left(\mathrm{Equation}9\right)\\ \mathrm{R3}=\frac{\begin{array}{c}\left[\mathrm{X4}+\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]\\ \left[\mathrm{R4}\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]-\\ \left[\mathrm{X4}\mathrm{R4}\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]\end{array}}{\left({\mathrm{R4}}^2+{\left[\mathrm{X4}+\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]}^2\right)}& \left(\mathrm{Equation}10\right)\end{array}$ $\begin{array}{cc}\mathrm{X3}=\frac{\begin{array}{c}\left[\mathrm{X4}\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]\\ \left[\mathrm{X4}+\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]+\\ \left[{\mathrm{R4}}^2\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]\end{array}}{{\mathrm{R4}}^2+{\left[\mathrm{X4}+\mathrm{Z52}\tan \left(52\mathrm{D52}\right)\right]}^2}& \left(\mathrm{Equation}11\right)\end{array}$

[0084] The equivalent impedance (ZI) for the driver circuit (20) can be derived from the circuit model of FIG. 2A, and is given by Equation 12. The resistive component (RI) of the input impedance (ZI) to the driver side of the matching network (50) is given by Equation 13, and the reactive component (XI) of the input impedance (ZI) to the driver side of the matching network (50) is given by Equation 14. Equations 13 and 14 can be derived for a parallel driver circuit capacitance (C20) and driver circuit resistance (R20) in series with a driver inductance (L32), as shown in FIG. 2A. $\begin{array}{cc}\mathrm{ZI}=\frac{\mathrm{R20}\left(1/j\mathrm{C20}\right)}{\mathrm{R20}+\left(1/j\mathrm{C20}\right)}+j\mathrm{L32}& \left(\mathrm{Equation}12\right)\end{array}$ $\begin{array}{cc}\mathrm{RI}=\frac{\mathrm{R20}}{1+{\left(\mathrm{R20}\mathrm{C20}\right)}^2}& \left(\mathrm{Equation}13\right)\end{array}$ $\begin{array}{cc}\mathrm{XI}=\frac{-\left({\mathrm{R20}}^2\mathrm{C20}\right)}{1+{\left(\mathrm{R20}\mathrm{C20}\right)}^2}+\mathrm{L32}& \left(\mathrm{Equation}14\right)\end{array}$

[0085] Referring again to FIG. 5, for impedance matching, the reactive components of the net input impedance (Z3) at the driver side of the matching network (50) and of the equivalent impedance (ZI) for the driver circuit (20) must cancel out (i.e., the matching network (50) must provide a reactance, which in combination with the driver and receiver circuits, will give no net reactance). To do this, the reactive component (X3) must be the conjugate (equal in magnitude with the opposite sign) of the reactive component (XI), as shown in Equation 15. The matching of the reactive components will result in no overshoots or undershoots.

[0086] Equations 11 and 14 can be substituted into Equation 15. Selected values for frequency (f) and length (D52) of the adjustable-length transmission line (52) can be substituted into Equation 15 to determine, by the process of iteration, the values at which the net reactance (XIX3) is equal to zero (0), thereby achieving impedance matching. FIG. 7 shows the values for length (D52) of the adjustable-length transmission line (52) as a function of frequency (f) which provide impedance matching over a range of selected values for the receiver circuit resistance (termination load) (R10) ranging from 50 ohms to 500 mega ohms. The values shown in FIG. 7 are determined using typical values as previously provided for C10 (3.7 pF), L31 (2 nH), D40 (76 mm or 3 inches), Z40 (50 ohms), 40 (4.5), L32 (2 nH), C20 (4.7 pF), Z52 (50 ohms), and 52 (4.5).

[0087] Additional iterative solutions were plotted for the length (D52) of the adjustable-length transmission line (52) as a function of frequency (f) over a range of selected values for L31, L32, C10, C20, 40, 52, D40, Z40, and Z52, successively. FIG. 8 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a receiver inductance (L31) in the range between 1 nH and 8 nH. The values shown in FIG. 8 are determined using typical values as previously provided for C10 (3.7 pF), R10 (50 ohms), D40 (76 mm or 3 inches), Z40 (50 ohms), 40 (4.5), L32 (2 nH), C20 (4.7 pF), Z52 (50 ohms), and 52 (4.5).

[0088] FIG. 9 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a driver inductance (L32) in the range between 1 nH and 16 nH. The values shown in FIG. 9 are determined using typical values as previously provided for C10 (3.7 pF), R10 (50 ohms), D40 (76 mm or 3 inches), Z40 (50 ohms), 40 (4.5), L31 (2 nH), C20 (4.7 pF), Z52 (50 ohms), and 52 (4.5).

[0089] FIG. 10 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a receiver capacitance (C10) in the range between 1 pF and 12 pF. The values shown in FIG. 10 are determined using typical values as previously provided for C20 (4.7 pF), R10 (50 ohms), D40 (76 mm or 3 inches), Z40 (50 ohms), 40 (4.5), L31 (2 nH), L32 (2 nH), Z52 (50 ohms), and 52 (4.5).

[0090] FIG. 11 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a driver capacitance (C20) in the range between 1 pF and 12 pF. The values shown in FIG. 11 are determined using typical values as previously provided for C10 (3.7 pF), R10 (50 ohms), D40 (76 mm or 3 inches), Z40 (50 ohms), 40 (4.5), L31 (2 nH), L32 (2 nH), Z52 (50 ohms), and 52 (4.5).

[0091] FIG. 12 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a dielectric constant (40, 52) for the transmission line (40) and the adjustable-length transmission line (52) in the range between 2 and 6. The values shown in FIG. 12 are determined using typical values as previously provided for C10 (3.7 pF), C20 (4.7 pF), R10 (50 ohms), D40 (76 mm or 3 inches), Z40 (50 ohms), L31 (2 nH), L32 (2 nH), and Z52 (50 ohms).

[0092] FIG. 13 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a length (D40) of the transmission line (40) in the range between 51 mm (2 inches) and 127 mm (5 inches). The values shown in FIG. 13 are determined using typical values as previously provided for C10 (3.7 pF), C20 (4.7 pF), R10 (50 ohms), Z40 (50 ohms), Z52 (50 ohms), 40 (4.5), L31 (2 nH), L32 (2 nH), and 52 (4.5).

[0093] FIG. 14 shows that the present invention can provide impedance matching using a comparable length (D52) adjustable-length transmission line (52) for a characteristic impedance (Z40, Z52) of the transmission line (40) and of the adjustable-length transmission line (52) in the range between 20 ohms and 200 ohms. The values shown in FIG. 14 are determined using typical values as previously provided for C10 (3.7 pF), C20 (4.7 pF), R10 (50 ohms), D40 (76 mm or 3 inches), 40 (4.5), L31 (2 nH), L32 (2 nH), and 52 (4.5).

[0094] Although illustrated and described above with reference to certain specific embodiments, the present invention is nevertheless not intended to be limited to the details shown. Rather, various modifications may be made in the details within the scope and range of equivalents of the claims and without departing from the spirit of the invention.