NPF radiofrequency device

[0001] This application claims priority to foreign French patent application No. FR 2315348, filed on Dec. 27, 2023, the disclosure of which is incorporated by reference in its entirety.

[0002] The invention lies in the field of differential IQ generator devices (or differential IQ combiners) at radiofrequency (RF) frequencies, that is to say for example 0.1-5 GHz or above.

[0003] As is known and illustrated in the left-hand part of FIG. 11, the typical function of an IQ generator is as follows: an IQ generator receives an input signal, V_RF,ant and makes it possible to obtain, at output, two quadrature signals (complex signals, providing a complex representation of the input signal), referred to as I⁺ and Q⁺, of the same frequency and phase shifted by 90°: I⁺=V_RF,out,0° and Q⁺=V_RF,out,90° (typical generator 81) or four signals in the case of a differential IQ generator with, in addition, I⁻=−V_RF,out,0°=V_RF,out,180° and Q⁻=−V_RF,out,90°=V_RF,out,270° (typical generator 81′).

[0004] Reciprocally, the function of such a device may be reversed, by exchanging one or more inputs and outputs: this then gives a typical IQ combiner 82 (possibly typical differential IQ combiner 82′) as illustrated in the right-hand part of FIG. 11. Such an IQ combiner receives, as inputs, two quadrature signals of the same frequency and phase shifted by 90° (complex signals): I⁺=V_RF,in,0° and Q⁺=V_RF,in,90° (combiner 82) or four signals in the case of the differential IQ combiner with, in addition, I″=−V_RF,in,0°=V_RF,in,180° and Q⁻=−V_RF,in,90°=V_RF,in,270° (combiner 82′), combines them and delivers the signal V_RF,out.

[0005] J. Kaukovuori, K. Stadius, J. Ryynanen, and K. A. I. Halonen, “Analysis and Design of Passive Polyphase Filters”, IEEE Transactions on Circuits and Systems I: Regular Papers, vol. 55, no 10, p. 3023-3307, November 2008, doi: 10.1109/TCSI.2008.917990, proposes, as shown in FIG. 1, two polyphase filters, PPF-I and PPF-II, for generating differential I and Q quadrature signals (I⁺, I⁻, Q⁺, Q⁻). Each of the filters has 4 paths.

[0006] However, these PPF architectures have the following drawbacks:

• • [0007] they have little or no frequency selectivity of the processed radiofrequency signal, thereby making the associated receivers sensitive to blockers in the case where the input signal is the signal received by an RF receiver;
  • [0008] they are not frequency agile;
  • [0009] they are bulky;
  • [0010] they theoretically exhibit at least 3 dB of losses between the RF input and the I, Q outputs.

[0011] There is therefore a need to have IQ generators (or IQ combiners), notably differential ones, at RF frequencies (0.1-5 GHz and above) that do not exhibit these drawbacks.

[0012] To this end, according to a first aspect, the present invention describes a radiofrequency IQ quadrature signal generator device comprising a first end terminal, four first branches called BR1₀, BR1₁, BR1₂, BR1₃ connected to said first end terminal,

• • [0013] each first branch comprises a capacitor, respectively C₀, C₁, C₂, C₃, having the same capacitance, comprising a first terminal and a second terminal, said second terminal of the capacitor being connected to ground and a switch arranged between the first end terminal and the first terminal of the capacitor of said first branch;
  • [0014] said device being designed to, in each first branch BR1_i, close the switch at each time pT₀+(i)T₀/4, for a time interval T₀/4, where p is a natural number and T₀ is the period of the clock signal;
  • [0015] said device being characterized in that:
  • [0016] the NPF device comprises at least one pair of second branches comprising two successive second branches in a cyclic ordered list of four second branches BR2₀, BR2₁, BR2₂, BR2₃, the second branch BR2_k comprising a second end terminal E2_k, and
  • [0017] each second branch BR2_k of the device comprises, for each first branch, BR1_i, i=0 to 3, a switch between the first terminal of the capacitor C_i of said first branch and said second branch, said device being designed to close said switch at each time pT₀+(i+k)T₀/4, for a time interval T₀/4;
  • [0018] said radiofrequency IQ quadrature signal generator device being designed to receive, at input, an input signal of frequency f_RF on said first end terminal, to generate at least two quadrature signals, of frequency f_RF, from the input signal and to deliver them, at output, on the second end terminals of said pair of second branches;
  • [0019] said second end terminals of the second branches of said pair being distinct from said first end terminal,
  • [0020] the frequencies f_RF and 1/T₀ being substantially equal;
  • [0021] for each second terminal of the device: the signal delivered at output on said second terminal of the device is linked to the input signal by a transfer function comprising a filter function and a phase shift function.

[0022] The invention is thus based on the use of a base of what are known as N-path filters (NPF), corresponding to N parallel capacitors switched by clock signals with a duty cycle 1/N. The NPF architecture generates (or conversely combines) N amplitude-balanced signals, with a period T₀, delayed with respect to one another by T₀/N. The architecture reduces losses: approximately 1.8 dB per path (compared to 3 dB of losses per path in PPF-I and PPF-II filters) and naturally offers frequency filtering with a high quality factor.

[0023] It does not require any resistors or inductors, thereby reducing bulk compared to phase shifters from the prior art.

[0024] In some embodiments, such a device comprises four second branches BR2₀, BR2₁, BR2₂, BR2₃ each comprising a second end terminal E2₀, E2₁, E2₂ and E2₃, respectively, and

• • [0025] each second branch BR2_k, k=0 to 3, comprises, for each first branch, BR1_i, i=0 to 3, a switch between the first terminal of the capacitor C_i of said first branch and said second branch, said device being designed to close said switch at each time pT₀+(i+k)T₀/4, for a time interval T₀/4,
  • [0026] said device being designed to receive an input signal of frequency f_RF on said first end terminal (E1) and to generate four quadrature signals in differential form, I⁺, I⁻, Q⁺, Q⁻, delivered on the second end terminal E2₀, E2₂, E2₁, E2₃, each of frequency f_RF,
  • [0027] said second end terminals E2₀, E2₁, E2₂ and E2₃ being distinct from said first end terminal;
  • [0028] it furthermore comprises a complex filter, the transfer function of which is not symmetrical about zero, and at the input of which the signals delivered to the second end terminals are supplied at the input of said complex filter.

[0029] According to another aspect, the invention describes a method for generating IQ signals using said device according to the first aspect of the invention, said method comprising the following steps:

• • [0030] applying an input signal to said first end terminal,
  • [0031] obtaining at least two quadrature signals on the second end terminals of at least said two successive second branches.

[0032] In one embodiment, this generation method using said device according to the first aspect of the invention in the indicated configuration in which it comprises four second branches BR2₀, BR2₁, BR2₂, BR2₃ comprises the following steps:

• • [0033] applying an input signal to said first end terminal,
  • [0034] obtaining the differential signal I⁺, respectively I⁻, Q⁺, Q⁻, on the second end terminal E2₀, respectively E2₂, E2₁, E2₃.

[0035] According to another aspect, the invention proposes a method for combining quadrature IQ signals using the device according to the first aspect of the invention, said method comprising the following steps:

• • [0036] applying two quadrature signals, one to the second end terminal of one of the two successive second branches, the other to the second end terminal of the other of the two successive second branches;
  • [0037] obtaining a signal resulting from the combination of said quadrature signals on said first end terminal.

[0038] In one embodiment, this generation method using said device according to the first aspect of the invention in the indicated configuration in which it comprises four second branches BR2₀, BR2₁, BR2₂, BR2₃ comprises the following steps:

• • [0039] applying the differential signal I⁺, respectively I⁻, Q⁺, Q⁻, to the second end terminal E2₀, respectively E2₂, E2₁, E2₃
  • [0040] obtaining a signal resulting from the combination of said differential signals on said first end terminal.

[0041] According to another aspect, the invention proposes a radiofrequency IQ quadrature signal combiner device comprising a first end terminal, four first branches called BR1₀, BR1₁, BR1₂, BR1₃ connected to said first end terminal,

• • [0042] each first branch comprises a capacitor, respectively C₀, C₁, C₂, C₃, having the same capacitance, comprising a first terminal and a second terminal, said second terminal of the capacitor being connected to ground and a switch arranged between the first end terminal and the first terminal of the capacitor of said first branch;
  • [0043] said device being designed to, in each first branch BR1_i, close the switch at each time pT₀+ (i)T₀/4, for a time interval T₀/4, where p is a natural number and T₀ is the period of the clock signal;
  • [0044] said radiofrequency IQ quadrature signal combiner device being characterized in that:
  • [0045] the device is an NPF device and comprises at least one pair of second branches comprising two successive branches in a cyclic ordered list of second branches BR2₀, BR2₁, BR2₂, BR2₃, the second branch BR2_k comprising a second end terminal E2_k, and
  • [0046] each second branch BR2_k of the device comprises, for each first branch, BR1_i, i=0 to 3, a switch between the first terminal of the capacitor C_i of said first branch and said second branch, said device being designed to close said switch at each time pT₀+(i+k)T₀/4, for a time interval T₀/4;
  • [0047] said device being designed to receive, at input, at least two quadrature signals of frequency f_RF, one on the second end terminal of one of the second branches of said pair of second branches, the other on the second end terminal of the other of the second branches of said pair of second branches, to combine them and to deliver, at output, on said first end terminal, a signal of frequency f_RF resulting from the combination of said quadrature signals;
  • [0048] said second end terminals of the second branches of said pair being distinct from said first end terminal;
  • [0049] the frequencies f_RF and 1/T₀ being substantially equal;
  • [0050] each signal received at input on a second terminal of said pair being linked to the signal delivered at output on the first terminal by a transfer function comprising a filter function and a phase shift function.

[0051] In one embodiment, such a device comprises four second branches BR2₀, BR2₁, BR2₂, BR2₃ each comprising a second end terminal E2₀, E2₁, E2₂ and E2₃, respectively, and

• • [0052] each second branch BR2_k, k=0 to 3, comprises, for each first branch, BR1_i, i=0 to 3, a switch between the first terminal of the capacitor C_i of said first branch and said second branch, said device being designed to close said switch at each time pT₀+ (i+k)T₀/4, for a time interval T₀/4,
  • [0053] said device being designed to receive, at input, four quadrature signals in differential form, I⁺, I⁻, Q⁺, Q⁻, of frequency f_RF, on the second end terminals E2₀, E2₂, E2₁, E2₃, to combine them and to deliver, at output, a signal of frequency f_RF resulting from the combination of said differential signals on said first end terminal;
  • [0054] said second end terminals E2₀, E2₁, E2₂ and E2₃ being distinct from said first end terminal,
  • [0055] each signal received at input on a second terminal of said pair being linked to the signal delivered at output on the first terminal by a transfer function comprising a filter function and a phase shift function.

[0056] According to another aspect, the invention proposes a method for combining quadrature IQ signals using the radiofrequency IQ quadrature signal combiner device (10) according to the invention, said method comprising the following steps:

• • [0057] applying, at the input of the device, two quadrature signals of frequency f_RF, one to the second end terminal of one of the second branches of said pair, the other to the second end terminal of the other of the second branches of said pair;
  • [0058] obtaining, at the output of the device, a signal of frequency f_RF resulting from the combination of said quadrature signals on said first end terminal.

[0059] In one embodiment, such a method comprises the following steps:

• • [0060] applying, at the input of the device, four quadrature differential signals I⁺, I⁻, Q⁺, Q⁻ to the second end terminals E2₀, E2₂, E2₁, E2₃, each of said signals being of frequency f_RF;
  • [0061] obtaining, at output, on said first end terminal, a signal of frequency f_RF resulting from the combination of said differential signals.

[0076] Identical references may be used in various figures to designate identical or comparable elements.

[0077] Preamble regarding N-path filters NPF

[0078] In the preamble, a few general features regarding N-path filters NPF will first be presented.

[0079] FIGS. 2, 3, 4 and 5 relate to a conventional NPF architecture.

[0080] In these figures, the source signal, for example from an antenna, is called V_RF,ant and its source impedance is called Ra. The signal V_RF,in is the one located after the impedance Ra with respect to the antenna. The input impedance equivalent to the circuit is denoted Z_in. It will be seen in FIGS. 2 to 5 that V_RF,out=V_RF,in.

[0081] An NPF filter is an electronic circuit that uses N paths, which are embodied by an identical number N of capacitors so as to improve the rejection of unwanted signals. It is often used in RF to reduce the impact of interference due to blockers.

[0082] More specifically, FIG. 3 is a diagram of an N-path filter-N-path mixer architecture.

[0083] FIGS. 2, 3 and 4 illustrate the case N=4, with the signal V_RF,ant=A(t)cos(ω_RFt+φ(t)), A(t) and φ(t) modelling amplitude modulation and phase modulation, respectively. For the sake of simplicity, it is chosen hereinafter to adopt A(t)=A, which is constant, and φ(t)=0° in the representations; since modulations are slow phenomena, this does not impact the theory. @RF is the angular frequency of the radiofrequency signal; ω_RF=2πf_RF where f_RF is the frequency of the radiofrequency signal. The NPF filter consists in sampling, for a period T₀/N, the signal V_RF,in on each of the paths so as to charge the corresponding capacitor only during this time interval. The action is repeated on each of the N paths with a delay between the control signals for the samplers (switches, commutators) of T₀/N. If @RF is close to ω₀=2πf₀=2π/T₀, then the signal is in the passband of the filter, where f₀ is the fundamental frequency of the control signal of period T₀ and ω₀ is the corresponding angular frequency. Here, “close” is understood to mean when the distance between the values is less than half the passband of the N-path filter defined below (for example based on the transfer function defined in paragraph 114). T_on is the on time of the clock over one phase.

[0084] With respect to FIGS. 2 to 4: the control signal for the sampler intended to be supplied to the capacitor referenced C₀ is called φ₀; the control signal for the sampler intended to be supplied to the capacitor referenced C₁ is called φ₉₀; the control signal for the sampler intended to be supplied to the capacitor referenced C₂ is called ω₁₈₀; the control signal for the sampler intended to be supplied to the capacitor referenced C₃ is called ω₂₇₀.

[0085] Across the terminals of each capacitor C_i there is a voltage signal, V_Ci, said to be at intermediate frequency FI with an angular frequency ω_FI=|ω_RF−ω₀|. For this reason, NPF filters that naturally provide a mixing function may be used as N-path mixer (NPM) in the literature.

[0086] FIG. 2 is a timing diagram of the electrical signals (V_RF,in, V_RF,ant, and the respective control signals φ₀, φ₉₀, φ₁₈₀, φ₂₇₀ for the switches) implemented in an NPF of the type shown in FIG. 3, and FIG. 4 is a timing diagram of the signal V_{RF,out,φ=0°} in an NPF circuit of the type shown in FIG. 3. It may be clearly seen that V_{RF,out,φ=0°}=V_RF,in

[0087] The equivalent circuit diagram of such a circuit is shown in FIG. 5, differentiating the two functions inherent to the circuit of FIG. 3: filtering function (corresponding to the block FLTR 110) and mixing function (corresponding substantially to the block MX 120).

[0088] For the mixing function, a signal of angular frequency ω_FI and amplitude-modulated and phase-modulated by A(t) and φ(t) is found on the node V_BB. V_BB indifferently represents (that is to say whatever the value of i) the voltage V_Ci across the terminals of the capacitor C_i.

[0089] For the filtering function, the ratio

[0090] $A_{V,\mathrm{RF}}\left(f_{\mathrm{RF}}\right)=\frac{V_{\mathrm{RF},\mathrm{out}}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)}$
is what is of interest.

[0091] The data in the equivalent circuit diagram at the frequency f_RF close to the clock frequency of the switches are as follows:

• • [0092] R_B=γ_NR_L, the dynamic resistance equivalent to the N parallel paths,
  • [0093] with R_L the low-frequency load resistance (example: that of a measuring amplifier),
  • [0094] where

[0095] ${\gamma }_N=\frac{1}{N}\sin c^2\left(\frac{\pi }{N}\right)$

• • [0096] Z_in is the input impedance of the circuit at the RF frequency seen from the antenna

[0097] $Z_{\mathrm{in}}=R_{\mathrm{sw}}+R_BR_{\mathrm{sh}}(\mathrm{signifies}{ }^{‶}\mathrm{in}{\mathrm{parallel}}^{″})$

• • [0098] where R_sh represents the power losses associated with R_α and R_sw due to up-conversion and down-conversion of signal harmonics

[0099] $R_{\mathrm{sh}}={\alpha }_N\left(R_a+R_{\mathrm{sw}}\right)$

• • [0100] where

[0101] ${\alpha }_N=\frac{N·{\gamma }_N}{1-N·{\gamma }_N}$

• • [0102] with R_sw the resistance of the switch and R_α the resistance of the source (or antenna resistance)

[0103] $C_B=\frac{C_i}{2{\gamma }_N}=\frac{C_L}{2{\gamma }_N},$
the dynamic capacitance equivalent to the N capacitors C_i in parallel. Whatever the value of i, C_i=C_L such that 2R_BC_B=R_L C_L

[0104] $L_B=\frac{1}{{\left(2\pi f_0\right)}^2{C}_B},$
the equivalent dynamic inductance of the filtering function

[0105] $V_{\mathrm{Ci}}=A\left(t\right)\cos \left(\phi \left(t\right)+\frac{2\pi \left(i-1\right)}{N}\right)\sin c\left(\frac{\pi }{N}\right),$
the voltage across the terminals of the capacitor C_i, φ(t) and A(t) being, as a reminder, slowly variable modulation functions. (FIG. 3).

[0106] This then results in the following amplitude filtering function, which naturally has a high quality factor:

[0107] $❘A_{V,\mathrm{RF}}\left(f_{\mathrm{RF}}\right)❘=❘\frac{V_{\mathrm{RF},\mathrm{out}}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)}❘=❘\frac{Z_{\mathrm{in}}}{Z_{\mathrm{in}}+R_a}❘=❘\frac{Z_B+R_{\mathrm{sw}}}{Z_B+R_{\mathrm{sw}}+R_a}❘$ $\mathrm{with}{Z}_B=\frac{j{\omega }_{\mathrm{RF}}{L}_B}{1+{\left(j{\omega }_{\mathrm{RF}}\right)}^2{L}_B{C}_B+j{\omega }_{\mathrm{RF}}{L}_B/\left(R_{\mathrm{sh}}R_B\right)}$

[0108] One particular case is that where R_L is chosen so as to have matching at input:

[0109] $R_L=R_{L\_\mathrm{match}}=R_a\frac{1}{{\gamma }_N}\frac{{\alpha }_N·\left(1+\rho \right)\left(1-\rho \right)}{{\alpha }_N·\left(1+\rho \right)-\left(1-\rho \right)}\mathrm{with}\rho =\frac{R_{\mathrm{sw}}}{R_a}.$

[0110] The NPF-based structure will always have its filtering characteristic with a particular value for

[0111] $R_B=R_{\mathrm{B\_match}}=\frac{R_{\mathrm{sh}}\left(R_a+R_{\mathrm{sw}}\right)}{R_{\mathrm{sh}}-\left(R_a+R_{\mathrm{sw}}\right)}.$

[0112] The invention lies in the context of N-path filters. It proposes an NPF architecture generating 4 signals that are sampled, of period T₀, delayed with respect to one another by T₀/4, and amplitude-balanced.

[0113] Reciprocally, the invention also makes it possible to address the inverse problem, namely to combine 4 periodic signals of period T₀ that are phase-shifted with respect to one another by T₀/4, and amplitude-balanced.

[0114] FIG. 6 shows an architecture of a 4-path NPF device 10 in one embodiment of the invention, which makes it possible to generate and combine differential IQ signals.

[0115] The NPF device 10 comprises a first end terminal, E1, and four first parallel branches, BR1₀, BR1₁, BR1₂, BR1₃, each connected to E1. The input voltage of the device, V_RF,in, is applied to E1 and therefore across the terminals of each of these four branches.

[0116] Each of these branches BR1_i comprises a switch (or commutator) 20 and a capacitor C_i, of value C_L (identical for the four capacitors), i=0 to 3: one terminal of this switch 20 is brought to the voltage V_RF,in, and the other terminal of the switch is connected to a first terminal of this capacitor. The second terminal of the capacitor is connected to ground.

[0117] The switch 20 connected to the capacitor C_i, i=0 to 3, is controlled by a control signal called φ_90x(i) (i.e. φ₀ for i=0, φ₉₀ for i=1, φ₁₈₀ for i=2, φ₂₇₀ for i=3) that causes it to close at each time pT₀+ (i)T₀/4 and to do so for a time interval T₀/4, where p is a natural number varying from 0 to P, the switch being open otherwise. P×T₀ is the operating time of the device, which may reach for example several minutes (more than Feb. 5, 2010 min for example) and T₀ adopts variable values, for example notably in a range of 0.04 to 10 μs.

[0118] The NPF device 10 therefore has 4 paths: indeed, it comprises four capacitors C_i, i=0 to 3.

[0119] Each capacitor C_i is thus selectively charged by the input signal of the device, V_RF,in, at the time pT₀+ (i)T₀/4, for a time interval T₀/4, by way of the respective switch 20 arranged upstream of each capacitor and driven by the control signal π_90x(i).

[0120] The NPF device 10 comprises four second parallel branches, BR2_k, k=0 to 3. The resistor RL represents the load of the circuit placed at the output of the device 10.

[0121] The voltage V_Ci across the terminals of C_i, i=0 to 3, is applied successively across the terminals of each of these four branches BR2₀, BR2₁, BR2₂, BR2₃ for a time interval T₀/4, at specific times, by way of respective switches 20 driven by the control signals φ_90x(i), as described below.

[0122] The voltage, V_Ci, across the terminals of the capacitor C_i is thus selectively applied to the branch BR2_k at each time pT₀+(i+k)T₀/4, for a time interval T₀/4. The corresponding control signals are indicated at the switches in FIG. 6.

[0123] The value of the voltage across the terminals of C_i is read 4 times, during a time interval T₀/4, every pT₀+k T₀/4 where k∈[0; 3], starting, for the branch BR2₀, at the time pT₀+(i)T₀/4.

The device 10 comprises 4 second end terminals, E2₀, E2₁, E2₂ and E2₃.

[0124] V_{RF,out,φ=0°} is the voltage signal on the branch BR2₀ (across the terminals of the resistor RL of this branch, at E2₀);

[0125] V_{RF,out,φ=90°} is the voltage signal on the branch BR2₁ (across the terminals of the resistor RL of this branch, at E2₁);

[0126] V_{RF,out,φ=180°} is the voltage signal on the branch BR2₂ (across the terminals of the resistor RL of this branch, at E2₂);

[0127] V_{RF,out,φ=270°} is the voltage signal on the branch BR2₃ (across the terminals of the resistor RL of this branch, at E2₃);

[0128] The phase called V_{RF,out,φ=2kπ/4=360° k/4} is the version delayed by kT₀/4 (phase delay of 360° K/4) of the signal V_RF,in, that is to say the version delayed by kT₀/4, and filtered and sampled, of the signal V_RF,ant.

[0129] The timing diagrams of the voltage V_{RF,out,φ=360°k/4} are shown opposite the point E2_k at this voltage in FIG. 6, for k=0 to 3.

[0130] The signal V_{RF,out,φ0=0°} is special in that it reproduces the signal V_RF,in without a delay because the switch for transferring charge from the capacitor to R_L is open at the same time as the switch for charging the capacitor. However, since the discharge time R_L. C_i=R_L. C_L is very long compared to T₀ (for example more than 10 times longer), discharging the capacitor at the same time as charging it has an impact similar to discharging it for an identical duration of T₀/4 but without charging it during this time, this being the case for the other three signals. This has resulted in the generation of 4 signals of period T₀ that are sampled, delayed with respect to one another by T₀/4, and naturally amplitude-balanced.

[0131] The invention makes it possible to address the following specific problem: the generation of 2 signals phase-shifted by 90° (IQ generator), as shown in FIG. 11 in the form of differential signals I+, I+, Q+, Q− where:

[0132] $\begin{array}{c}{I}^{+}=V_{\mathrm{RF},\mathrm{out},\varphi =0°}\\ I^{-}=V_{\mathrm{RF},\mathrm{out},\varphi =180°}\\ Q^{+}=V_{\mathrm{RF},\mathrm{out},\varphi =90°}\\ Q^{-}=V_{\mathrm{RF},\mathrm{out},\varphi =270°}\end{array}$

[0133] In order to validate the model, to verify the equality of the amplitude of the voltage gains |A_V,RF,φk(f_RF)| for φ=0 (charge and discharge during the same cycle) and φ≠0 (charge and discharge during a different cycle), as well as the equal distribution of delays, two cases were studied as a function of capacitance C_i and are illustrated in FIGS. 8 and 9, for which the outputs V_RF,out,φk (f_RF) are loaded by a high impedance R_L=100 kΩ. The values of the parameters are explained in a key.

[0134] FIG. 8 is a frequency representation of 4 IQ signals generated in one embodiment of the invention, revealing the balancing of amplitudes and filtering properties: the 4 output phases are well balanced around the center frequency of the filter. As with NPFs, the filtering properties are linked to the value of the capacitor C_i=C_L.

[0135] FIG. 9 is a temporal representation of 2 differential IQ signals generated by the invention, showing the balancing of the delays on the timing diagrams: the 4 output phases indeed have a delay of

[0136] $\frac{T_0}{4}$
with respect to one another, corresponding to a phase shift of 90°. The following differential signals I and Q are thus found:

[0137] $\begin{array}{c}{I}^{+}=V_{\mathrm{RF},\mathrm{out},\varphi =0°}\\ I^{-}=V_{\mathrm{RF},\mathrm{out},\varphi =180°}\\ Q^{+}=V_{\mathrm{RF},\mathrm{out},\varphi =90°}\\ Q^{-}=V_{\mathrm{RF},\mathrm{out},\varphi =270°.}\end{array}$

[0138] The equivalent circuit diagram of such a circuit is shown in FIG. 7, differentiating the two filtering (block FLTR 210) and phase shift (block DEφ 220) functions inherent to the circuit of FIG. 6.

[0139] For the phase shift function, the ratio

[0140] $A_{V,\mathrm{RF},{\varphi }_k}\left(f_{\mathrm{RF}}\right)=\frac{V_{\mathrm{RF},\mathrm{out},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)}=\frac{V_{\mathrm{RF},\mathrm{out},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{in}}\left(f_{\mathrm{RF}}\right)}·\frac{V_{\mathrm{RF},\mathrm{in},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)},$
which naturally integrates the filtering properties of the NPF, is what is of interest.

[0141] The data in the equivalent circuit diagram are as follows:

• • [0142] R_B=γ_NR_L′, the resistance equivalent to the N (with N=4) phase-shifted paths in parallel.
  • [0143] where R_L′=R_L+R_sw is the sum of the low-frequency load resistance (example: that of a measuring amplifier) and the resistance R_sw of each of the switches located after the capacitor.

[0144] ${\gamma }_N=\frac{1}{N}{\mathrm{sinc}}^2\left(\frac{\pi }{N}\right)$

• • • [0145] where
  • [0146] Z_in is the input impedance of the circuit at the RF frequency seen from the antenna

[0147] $Z_{\mathrm{in}}=R_{\mathrm{sw}}+R_BR_{\mathrm{sh}}$

• • [0148] where R_sh represents the power losses associated with R_α and R_sw due to up-conversion and down-conversion of signal harmonics

[0149] $R_{\mathrm{sh}}={\alpha }_N\left(R_a+R_{\mathrm{sw}}\right)$

• • [0150] where

[0151] ${\alpha }_N=\frac{N·{\gamma }_N}{1-N·{\gamma }_N}$

• • [0152] with R_sw the resistance of the switch and R_α the resistance of the source (or antenna resistance)

[0153] $C_B=\frac{c_i}{2{\gamma }_N}=\frac{c_L}{2{\gamma }_N},$
the dynamic capacitance equivalent to the N capacitors C_i in parallel. Whatever the value of i, C_i=C_L such that 2R_BC_B=R_L′C_L

[0154] $L_B=\frac{1}{{\left(2\pi f_0\right)}^2{C}_B},$
the equivalent dynamic inductance of the filtering function

[0155] $V_{\mathrm{Ci}}=A\left(t\right)\cos \left(\phi \left(t\right)+\frac{2\pi \left(i-1\right)}{N}\right)\mathrm{sinc}\left(\frac{\pi }{N}\right),$
the voltage across the terminals of the capacitor C_i, φ(t) and A(t) being, as a reminder, slowly variable modulation functions (FIG. 6).

[0156] There is the following phase shift function ∀k∈[1; N=4], at the frequency f_RF close to the clock frequency of the switches (cf. above for what close is understood to mean), i.e. these frequencies are substantially equal:

[0157] $\begin{array}{c}❘A_{V,\mathrm{RF},{\varphi }_k}\left(f_{\mathrm{RF}}\right)❘=❘\frac{V_{\mathrm{RF},\mathrm{out},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)}❘=❘\frac{V_{\mathrm{RF},\mathrm{out},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{in}}\left(f_{\mathrm{RF}}\right)}❘·❘\frac{V_{\mathrm{RF},\mathrm{in},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)}❘\\ \mathrm{with}❘\frac{V_{\mathrm{RF},\mathrm{in},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left(f_{\mathrm{RF}}\right)}❘=❘\frac{Z_{\mathrm{in}}}{Z_{\mathrm{in}}+R_a}❘=❘\frac{Z_B+R_{\mathrm{sw}}}{Z_B+R_{\mathrm{sw}}+R_a}❘\\ \mathrm{and}❘\frac{V_{\mathrm{RF},\mathrm{out},{\varphi }_k}\left(f_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{in}}\left(f_{\mathrm{RF}}\right)}❘=❘\frac{R_L}{R_L+R_{\mathrm{sw}}}❘·❘\frac{Z_B}{Z_B+R_{\mathrm{sw}}}❘\\ \mathrm{and}{Z}_B=\frac{j{\omega }_{\mathrm{RF}}{L}_B}{1+{\left(j{\omega }_{\mathrm{RF}}\right)}^2{L}_B{C}_B+j{\omega }_{\mathrm{RF}}{L}_B/\left(R_{\mathrm{sh}}R_B\right)}\end{array}$

[0158] That is to say:

[0159] $❘A_{V,\mathrm{RF},{\varphi }_k}\left(f_{\mathrm{RF}}\right)❘❘\frac{V_{\mathrm{RF},\mathrm{out},{\varphi }_k}\left({\omega }_{\mathrm{RF}}\right)}{V_{\mathrm{RF},\mathrm{ant}}\left({\omega }_{\mathrm{RF}}\right)}❘=\frac{R_L}{R_L+R_{\mathrm{sw}}}·❘\frac{Z_B}{Z_B+R_{\mathrm{sw}}+R_a}❘$

[0160] Reciprocally, the architecture as shown in FIG. 6 makes it possible to address the inverse problem: the combination of 2 quadrature differential signals (FIG. 11). The corresponding signals are shown in FIG. 10 in a case where the 4 inputs are fed by 4 antennas, of impedance R_α=500. The output is high-impedance R_L=100 kΩ. The inputs are respectively:

• • [0161] I⁺ on the input V_RF,ant,0°
  • [0162] I⁻ on the input V_RF,ant,180°
  • [0163] Q⁺ on the input V_RF,ant,90°
  • [0164] Q⁻ on the input V_RF,ant,270°.

[0165] FIG. 12 illustrates the generation of IQ signals at RF frequency in one embodiment of the invention and the use of the I and Q signals to implement a complex filter 83 (cf. K. W. Martin, “Complex signal processing is not complex,” in IEEE Transactions on Circuits and Systems I: Regular Papers, vol. 51, no. 9, pp. 1823-1836, September 2004, doi: 10.1109/TCSI.2004.834522). The first block is an IQ signal generator implementing the typical IQ generator functions as illustrated for example by the generator 81 (or alternatively 81′) of FIG. 11, and implemented according to the invention. The output signals therefore may or may not be differential. The following block 83 is a complex filter the transfer function of which is not symmetrical about zero. It may be an N-path filter or a continuous-time filter (without a clock). The advantage over a real filter is that it is easier to independently control the center frequency and the bandwidth. In addition, a complex filter may exhibit better signal rejection than a real filter.

[0166] In an IQ generator according to the invention, regardless of the configuration selected (i.e. with two output branches or four output branches), the output signals delivered on the end terminals E2_i of the output branches depend on the input signal applied to the input terminal E1 and are of the same frequency as this input signal.

[0167] In an IQ generator according to the invention, regardless of the configuration selected (i.e. with two output branches or four output branches), the output terminals of the device are distinct (electrically) from the input terminal (or from the input terminals in the case of a differential input). In other words, an output terminal of the IQ generator cannot at the same time be an input terminal of the IQ generator.

[0168] Similarly, in an IQ combiner according to the invention, regardless of the configuration selected (i.e. with two input branches or with four input branches), an output terminal of the IQ combiner cannot at the same time be an input terminal of the IQ combiner.

[0169] It should be noted that, in a device according to the invention, in its IQ combiner function or in its IQ generator function, the RC constant of each of the load circuits is very large compared to T₀/4 (at least 10 times greater), considering here for example that the resistance of each switch of the assembly is R_sw:

[0170] $\begin{array}{c}-\left(R_a+R_{\mathrm{SW}}\right)*C_L>>T_0;\\ -\left(R_L+R_{\mathrm{SW}}\right)*C_L>>T_0;\end{array}$

[0171] In one embodiment, the input signal is a signal in differential form. In this case, with reference to FIG. 6, the device 10′, in one embodiment of the invention, comprises notably an additional input terminal E1′ and is supplemented in the manner shown in FIG. 13.

[0172] The advantages conferred by an NPF device according to the invention are notably:

• • [0173] the natural frequency selectivity of the processed radiofrequency signal, inherent due to the use of an architecture based on switched capacitors;
  • [0174] associated receivers that are relatively insensitive to blockers;
  • [0175] an architecture that is intrinsically tuneable over a wide frequency band (typically 0.1 GHz-5 GHz), without degrading performance on this band;
  • [0176] a more compact architecture;
  • [0177] an architecture that limits losses on each path (1.8 dB).