Oscillator circuit and current compensation circuit for oscillator

[0001] The present disclosure relates to oscillator circuits and, more particularly, to a current compensation circuit for an oscillator, and an oscillator circuit with immunity to power supply variations.

[0002] A system-on-chip (SoC) is an integrated circuit (IC) that incorporates components of an entire electronic system onto a single platform. Power supplies drive numerous functional blocks that are densely integrated on the same chip. The SoC can downsize what are normally multi-chip designs onto a single processor, thereby reducing energy waste and the space occupied by large systems. However, high operating frequencies of the SoC will cause the power supplies to drive rapidly changing loads, thus creating noise in the power supply buses. The created noise degrades the power supply rejection ratio (PSRR), and is propagated throughout the chip to create jitter, which deteriorates a dynamic range of a sensitive function block such as a phase-locked loop (PLL). In the advanced node SoC design, PSRR degradation is exacerbated due to the reduction in supply voltage. Thus, there is a need in the art for an improved design to reduce the adverse effects resulting from the power supply variations.

[0003] The described embodiments provide a current compensation circuit for an oscillator, and an oscillator circuit with high immunity to power supply variations.

[0004] Some embodiments described herein may include a current compensation circuit for an oscillator. The current compensation circuit includes a current generating circuit and a signal generator. The current generating circuit has a first node and a second node. The first node is coupled to an input signal. The second node is coupled to a process-dependent current. The current generating circuit is configured to provide a compensation current according to the input signal and the process-dependent current. The signal generator, coupled to the current generating circuit, is configured to couple a supply voltage to the first node to provide the input signal. A signal level of the input signal changes in response to a variation in the supply voltage. A supply node of an oscillator is arranged to receive a supply current to drive the oscillator. The current generating circuit is arranged to draw the compensation current from the supply node to compensate for a variation in the supply current generated in response to the variation in the supply voltage.

[0005] Some embodiments described herein may include an oscillator circuit. The oscillator circuit includes a current generator, and oscillator and a current compensation circuit. The current generator is supplied by a supply voltage to output a supply current. The oscillator has a supply node arranged to receive the supply current. The oscillator is arranged to generate an oscillating signal according to a driving current. A first portion of the supply current flows into the oscillator from the supply node of the oscillator to serve as the driving current. The current compensation circuit, coupled to the supply node of the oscillator, is configured to generate a first portion of a compensation current according to the supply voltage and a reference signal, and generate a second portion of the compensation current according to a first process-dependent current. A signal level of the reference signal is independent of a variation in the supply voltage, and the first portion of the compensation current changes in response to the variation in the supply voltage. The current compensation circuit is arranged to draw the compensation current from the supply node of the oscillator to compensate for a variation in the supply current generated in response to the variation in the supply voltage.

[0006] With the use of the proposed current compensation scheme, an oscillator circuit can generate an oscillating signal of a stable or substantially fixed frequency under power supply variations. The proposed current compensation scheme can enable the oscillator circuit to enhance the immunity to power supply noise. In addition, the oscillator circuit can be utilized to implement a PLL with reduced jitter and phase noise. The proposed current compensation scheme can improve the PSRR performance without increasing the voltage headroom of the PLL.

[0020] The following disclosure provides many different embodiments, or examples, for implementing different features of the provided subject matter. Specific examples of components and arrangements are described below to simplify the present disclosure. These are, of course, merely examples and are not intended to be limiting. For example, the present disclosure may repeat reference numerals and/or letters in the various examples. This repetition is for the purpose of simplicity and clarity and does not in itself dictate a relationship between the various embodiments and/or configurations discussed.

[0021] Further, it will be understood that when an element is referred to as being “connected to” or “coupled to” another element, it may be directly connected to or coupled to the other element, or intervening elements may be present.

[0022] To enhance the ability to reject power supply ripples, a PLL may utilize a high impedance path to reduce variations in a supply current injected to a current-controlled oscillator. For example, a cascode circuit that is implemented using two transistors connected in series can be provided between a power supply and the current-controlled oscillator in order to reduce ripples in the supply current. In some fabrication processes, however, the transistors employed in the cascode circuit exhibit relatively low output resistance. Connecting the cascode circuit between the power supply and the current-controlled oscillator is not enough to provide good jitter performance.

[0023] One method is to add a gain boosting stage at the output of the cascode circuit, thereby increasing the output impedance looking into the output of the cascode circuit. However, one or more transistors of the gain boosting stage will be connected in series with the transistors of the cascode circuit, resulting in an increase in voltage headroom. This method would not work for circuit design at advanced technology nodes.

[0024] The present disclosure describes exemplary current compensation circuits, each of which can generate a compensation current by emulating/simulating the effect of process, voltage and/or temperature variations on a supply current injected to a supply node of an oscillator. The current compensation circuit can draw the compensation current from the supply node to compensate for a variation in the supply current. For example, the compensation current can be sourced from the supply current, and can be equivalent to a portion of the supply current that changes in response to the process, voltage and/or temperature variations. Another portion of the supply current, which is unaffected or substantially unaffected by the process, voltage and/or temperature variations, can flow into the oscillator through the supply node to serve as a driving current for controlling an oscillation frequency of the oscillator. Thus, the oscillation frequency can be unaffected or substantially unaffected by power supply variations.

[0025] The present disclosure further describes exemplary oscillator circuits with high immunity to power supply variations. Each oscillator circuit can be employed in a PLL for reducing the phase noise and jitter. Further description is provided below.

[0026] FIG. 1 is a diagram illustrating an exemplary oscillator circuit in accordance with some embodiments of the present disclosure. The oscillator circuit 100 can be utilized to implement a PLL with high immunity to power supply noise. However, this is not intended to be limiting. Those skilled in the art will appreciate that the oscillator circuit 100 can be employed in other circuitry or devices without departing from the scope of the present disclosure. In addition, an oscillator included in the oscillator circuit 100 may be implemented using a current-controlled oscillator, in which an oscillation frequency is controlled by a control current injected thereto. This example is provided for illustrative purposes and is not to be construed as limiting. The proposed current compensation scheme can be applied other types of oscillators, such as a voltage-controlled oscillator or a hybrid controlled oscillator, without departing from the scope of the present disclosure.

[0027] The oscillator circuit 100 may include, but is not limited to, a current generator 102, an oscillator 104 and a current compensation circuit 110. The current generator 102 is supplied by a supply voltage VCC to output a supply current I_S. The oscillator 104 can be arranged to generate an oscillating signal CK_OSC according to a control current injected thereto. For example, the oscillator 104 has a supply node N_OSC that is arranged to receive the supply current I_S. The supply current I_S can flow into the supply node N_OSC to drive the oscillator 104. A driving current I_D sourced from the supply current I_S can serve as the control current of the oscillator 104. For example, a frequency of the oscillating signal CK_OSC can be controlled by the driving current I_D.

[0028] The current compensation circuit 110, coupled to the supply node N_OSC, can be arranged to draw a current I_C from the supply node N_OSC to thereby compensate for the effect of power supply noise on the supply current I_S. For example, a variation in the supply voltage VCC can cause a variation in the supply current I_S. The current I_C can be a compensation current generated by emulating/simulating the effect of power supply noise on the supply current I_S. The driving current I_D may stay at a constant or substantially constant level under power supply variations. Thus, the frequency of the oscillating signal CK_OSC may be kept at constant or substantially constant under power supply variations.

[0029] By way of example but not limitation, the current generator 102 may be implemented using a transistor M_B biased with a voltage V_B, and the oscillator 104 may be implemented using a ring oscillator. The oscillator 104 in combination with the current generator 102 can be regarded as a voltage-controlled oscillator. In the example of FIG. 1, the oscillator 104 may include N inverters A1-AN connected in series. N is an integer greater than one. Note that the current generator 102 and/or the oscillator 104 can be implemented using other circuit structures without departing from the scope of the present disclosure.

[0030] When the supply voltage VCC increases because of power supply noise, the supply current I_S will increase accordingly. The current compensation circuit 110 can generate the current I_C (e.g. a compensation current) which can exhibit the effect of power supply noise on the supply current I_S. The current I_C can be a portion of the supply current I_S that flows from the supply node N_OSC to the current compensation circuit 110. Another portion of the supply current I_S that flows from the supply node N_OSC to the oscillator 104 may serve as the driving current I_D. Note that the driving current I_D can be a result of subtraction of the current I_C from the supply current I_S, and therefore can stay at a constant or substantially constant level. By emulating/simulating the effect of power supply noise on the supply current I_S, the current compensation circuit 110 can reduce/eliminate the impact of the power supply noise on the frequency stability of the oscillator 104.

[0031] In some embodiments, process and/or temperature variations in the current generator 102 and the oscillator 104 may result in different responses of the supply current I_S to power supply noise. The current compensation circuit 110 may further emulate/simulate the effect of process and/or temperature variations to generate the current I_C for noise compensation.

[0032] For example, consider a case where a first circuit and a second circuit are both implemented using the same structure formed by the current generator 102 and the oscillator 104. When subjected to the same power supply noise or voltage variation, the first and second circuits may exhibit different current responses due to process variations. Additionally or alternatively, the first/second circuit may exhibit different current responses when subjected to the same power supply noise or voltage variation at different temperatures. In other words, a variation in a supply current induced by power supply noise may differ depending on process and/or temperature variations. The proposed current compensation scheme can compensate for a variation in a supply current that is caused by process, voltage and/or temperature variations.

[0033] FIG. 2 is an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiment of the present disclosure. The current compensation circuit 110 includes, but is not limited to, a signal generator 120 and a current generating circuit 130. The signal generator 120 can be configured to couple the supply voltage VCC to a node N_M1 of the current generating circuit 130 to provide an input signal S_IN. The signal level of the input signal S_IN can change in response to a variation in the supply voltage VCC. In other words, the input signal S_IN can be related to voltage variations. In some examples, the input signal S_IN can be a current signal that can indicate the effect of the variation in the supply voltage VCC on the supply current I_S. In some other examples, the input signal S_IN can be a voltage signal that carries information on the variation in (or a noise component of) the supply voltage VCC.

[0034] The nodes N_M1 and N_M2 of the current generating circuit 130 can be coupled to the input signal S_IN and the process-dependent current I_PT, respectively. The current generating circuit 130 can be configured to provide a compensation current (i.e. the current I_C) according to the input signal S_IN and the process-dependent current I_PT. Note that the current I_C can flow out of the supply node N_OSC to compensate for a variation in the supply current I_S that is caused by process, voltage and/or temperature variations.

[0035] The process-dependent current I_PT can be related to process variations. For example, if the proposed current compensation scheme was not applied to the oscillator 104, the supply current I_S would exhibit changes in response to process variations in transistor(s). To compensate for process variations, the current generating circuit 130 can provide the process-dependent current I_PT by emulating/simulating the effect of process variations on the supply current I_S. In some embodiments, the process-dependent current I_PT may be further dependent on temperature. The process-dependent current I_PT can be steered toward or away from the current generating circuit 130 to thereby compensate for process and temperature variations.

[0036] In operation, the supply current I_S can be supplied to the supply node N_OSC to drive the oscillator 104. The signal generator 120 can provide the input signal S_IN that can exhibit changes in response to a variation in the supply voltage VCC. The process-dependent current I_PT that is related to process and/or temperature variations can flow though the node N_M2. The current generating circuit 130 can draw the current I_C from the supply node N_OSC according to the input signal S_IN and the process-dependent current I_PT, thereby compensating for a variation in the supply current I_S that is generated in response to a variation in the supply voltage VCC. Note that the variation in the supply current I_S generated in response to the variation in the supply voltage VCC might differ depending on process and/or temperature variations. With the use of the current I_C that is related to not only the of power supply noise on the supply current I_S, but also the effect of process and/or temperature variations on the supply current I_S, the current generating circuit 130 can compensate for a variation in the supply current I_S that is caused by process, voltage and/or temperature variations. The oscillator 104 can generate the oscillating signal CK_OSC according to the driving current I_D having a constant or substantially constant level.

[0037] With the use of the proposed current compensation scheme, an oscillator circuit can generate an oscillating signal of a stable or substantially fixed frequency under power supply variations. The proposed current compensation scheme can enable the oscillator circuit to enhance the immunity to power supply noise. In addition, the oscillator circuit can be utilized to implement a PLL with reduced jitter and phase noise.

[0038] To facilitate understanding of the present disclosure, some embodiments are given as follows for further description of the proposed current compensation scheme. Those skilled in the art should appreciate that other embodiments employing the circuit structure shown in FIG. 1 or FIG. 2 are also within the contemplated scope of the present disclosure.

[0039] FIG. 3 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The oscillator circuit 300 may include a current generator 302, an oscillator 304 and a current compensation circuit 310. The current generator 302 and the oscillator 304 can be implemented using the current generator 102 and the oscillator 104 shown in FIG. 1, respectively. Each of the current generator 302 and the oscillator 304 can be represented by a current source symbol. In addition, the current compensation circuit 310 can be an embodiment of the current compensation circuit 110 shown in FIG. 2. The current compensation circuit 310 may include a signal generator 320 and a current generating circuit 330, which can be respective embodiments of the signal generator 120 and the current generating circuit 130 shown in FIG. 2.

[0040] In the present embodiment, the input signal S_IN shown in FIG. 2 can be an input current I_IN flowing into the node N_M1. For example, the current generating circuit 330 can be configured to provide a node voltage V_N at the node N_M1. A voltage level of the node voltage V_N is independent or almost independent of a variation in the supply voltage VCC. The signal generator 320 has a supply node N_CG coupled to the supply voltage VCC. The signal generator 320 can be configured to provide the input current I_IN according to a difference between the supply voltage VCC and the node voltage V_N. The input current I_IN flows into the node N_M1 to serve as an embodiment of the input signal S_IN shown in FIG. 2.

[0041] The signal generator 320 can be implemented using a current source arranged for providing the input current I_IN. In the example of FIG. 3, the signal generator 320 can be implemented using a resistor current source, which includes a resistive element R1 coupled between the supply node N_CG and the node N_M1. The magnitude of the input current I_IN can be equal to or substantially equal to the difference between the supply voltage VCC and the node voltage V_N divided by the resistance of the resistive element R1.

[0042] The current generating circuit 330 may include, but is not limited to, a current mirror 332 and a current generator 336. The current mirror 332 is coupled to the signal generator 320 through the node N_M1, and has a node N_C1 coupled to the supply node N_OSC. The current mirror 332 is arranged to provide the node voltage V_N at the node N_M1, draw a first portion of the input current I_IN (i.e. a current I_M1) from the node N_M1, and mirror the first portion of the input current I_IN to provide the current I_C. The current I_C can flow into the current mirror 332 from the node N_C1.

[0043] By way of example but not limitation, the current mirror 332 may include an amplifier 334, and transistors M41 and M42. The amplifier 334 has an input terminal TI1, an input terminal TI2 and an output terminal TO. The input terminal TI1 is coupled to a reference voltage VREF. The input terminal TI2 is coupled to the node NM1, and arranged to provide the node voltage VN at the node NM1 according to the reference voltage VREF. The reference voltage VREF may be insensitive to or independent of process, voltage and temperature (PVT) variations. For example, the reference voltage VREF can be provided by a bandgap voltage reference. In addition, the node voltage VN may be equal to or substantially equal to the reference voltage VREF. In the example of FIG. 3, the amplifier 334 can be implemented using an operational amplifier that has a very large or almost infinite gain, thereby forming a virtual short circuit in which the voltages at the inverting and noninverting terminals are equal or substantially equal.

[0044] Respective control terminals of the transistors M₄₁ and M₄₂ are coupled to the output terminal T_O. A first connection terminal of the transistor M₄₁ is coupled to the node N_M1, and the current I_M1 flows into the transistor M₄₁ from the first connection terminal of the transistor M₄₁. A first connection terminal of the transistor M₄₂ is coupled to the node N_C1, and the current I_C flows into the transistor M₄₂ from the first connection terminal of the transistor M₄₂. In addition, respective second connection terminals of the transistors M₄₁ and M₄₂ can be coupled to a reference voltage VSS, such as a ground voltage.

[0045] In addition, the current generator 336 is coupled to the supply voltage VCC through the node N_M2, and has a node N_C2 coupled to the node N_M1. The current generator 336 can be configured to draw a second portion of the input current I_IN (i.e. a current I_M2) from the node N_M1 according to the process-dependent current I_PT. The current generator 336 may include, but is not limited to, a current mirror 338 and a resistive element R2. The current mirror 338 can be arranged to mirror the process-dependent current I_PT to draw the current I_M2 (i.e. the second portion of the input current I_IN) from the node N_M2. The current I_M2 can be regarded as a process-dependent current.

[0046] Both of the process-dependent current I_PT and the current I_M2 can be related to the effect of process variations on the supply current I_S. By way of example but not limitation, the current generator 302 or the oscillator 304 can be a MOS-based circuit, which is developed based on metal-oxide-semiconductor field-effect transistors (MOSFETs). The process-dependent current I_PT can flow through a transistor of the current mirror 338, and hence can be related to a current response of the transistor to process variations. The current I_M2, produced in response to the process-dependent current I_PT, can also be related to a current response of the transistor to process variations. Note that the transistor of the current mirror 338 may exhibit temperature dependent characteristics. The process-dependent current I_PT and the current I_M2 can be further related to a current response of the transistor to temperature variations.

[0047] In the example of FIG. 3, the current mirror 338 may include the transistors M₃₁ and M₃₂. A first connection terminal of the transistor M₃₁ is coupled to the control terminal of the transistor M₃₁, and arranged to receive the process-dependent current I_PT. A control terminal of the transistor M₃₂ is coupled to the control terminal of the transistor M₃₁. A first connection terminal of the transistor M₃₂ is coupled to the node N_C2, and arranged to receive the current I_M2. In other words, the current I_M2 leaving the node N_M1 can flow into the transistor M₃₂ from the first connection terminal of the transistor M₃₂. In addition, respective second connection terminals of the transistors M₃₁ and M₃₂ are coupled to the reference voltage VSS. Note that each of the transistors M₃₁ and M₃₂ can be implemented as an n-channel transistor. The current I_M2 can be related to the effect of process variations in the oscillator 304 (or the current generator 302) if the oscillator 304 (or the current generator 302) is NMOS (n-channel MOSFET) dominant.

[0048] A terminal of the resistive element R2 is coupled to the supply voltage VCC through the node N_M2, and another terminal of the resistive element R2 is coupled to the first connection terminal of the transistor M₃₁. The process-dependent current I_PT can flow through a transistor of the current mirror 338 (e.g. the transistor M₃₁) and the resistive element R2. Note that the resistance of the resistive element R2 can change with temperature. Steering the process-dependent current I_PT to or from the resistive element R2 can achieve better temperature compensation. For example, the resistive element R2 can alleviate or prevent overcompensation of temperature variations.

[0049] In operation, the current generator 302 outputs the supply current I_S to the supply node N_OSC according to the supply voltage VCC. The signal generator 320 can provide the input current I_IN that varies with the supply voltage VCC. The input current I_IN can be divided into different portions at the node N_M1 that has a constant or substantially constant voltage, i.e. the node voltage V_N.

[0050] For example, the voltage difference across the resistive element R1 causes the input current I_IN to flow into the node N_M1. The voltage difference across the resistive element R2 causes the process-dependent current I_PT to flow into the node N_M2. The process-dependent current I_PT can be process and temperature dependent since the process-dependent current I_PT flows through the resistive element R2 and the transistor M₃₁. The current mirror 338 can draw the process-dependent current I_PT into the transistor M₃₁, and mirror the process-dependent current I_PT to provide the current I_M2, which is a portion of the input current I_IN that flows into the transistor M₃₂. Another portion of the input current I_IN flows into the transistor M₄₁ to serve as the current I_M1, which can be related to the effect of PVT variations.

[0051] In addition, the current mirror 332 can mirror the current I_M1 to provide the current I_C, i.e. a compensation current. The current I_C can flow out of the supply node N_OSC to compensate for a variation in the supply current I_S. The oscillator 304 can generate the oscillating signal CK_OSC of a stable or substantially fixed frequency according to the driving current I_D.

[0052] Note that as the process-dependent current I_PT can be related to a current response of a transistor to process variations, the oscillator circuit 300 can achieve a high PSRR over different process corners. In addition, the oscillator circuit 300 can have a low temperature coefficient with the aid of a temperature response of the process-dependent current I_PT. For example, when the supply current I_S increases in response to an increase in temperature, the process-dependent current I_PT flowing through the resistive element R2 and the transistor M₃₁ can decrease. The current I_M2 may decrease with a decrease in the process-dependent current I_PT, and the current I_M1 may increase with a decrease in the current I_M2, resulting in an increase in the current I_C. Thus, the driving current I_D may be kept at a substantially constant level. In other words, the oscillator circuit 300 can have a low temperature coefficient.

[0053] The circuit structure shown in FIG. 3 is provided for illustrative purposes, and is not intended to limit the scope of the present disclosure. In some embodiments, the node voltage V_N may be provided by other circuits or devices that are capable of generating a voltage independent of a variation in the supply voltage VCC. In some embodiments, the signal generator 320 can be implemented using other circuit structures, each of which can output a current that varies with the supply voltage VCC. In some embodiments, other circuits/devices capable of generating a temperature-dependent current can be used to replace the resistive element R2.

[0054] FIG. 4 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The oscillator circuit 400 may include the current generator 302 and the oscillator 304 shown in FIG. 3, and further include a current compensation circuit 410. The structure of the current compensation circuit 410 is similar/identical to that of the current compensation circuit 310 shown in FIG. 3 except for the current generator 436 included in the current generating circuit 430.

[0055] The current generator 436 may include the current mirror 338 shown in FIG. 3, a current mirror 438 and a resistive element R3. The current mirror 438 is coupled to the current mirror 338 through the node N_M3. The current mirror 438, coupled to the node N_M2, can be arranged to mirror the process-dependent current I_PT to provide an intermediate current (i.e. a current I_M3) flowing through a node N_M3. The current mirror 338, coupled to the current mirror 438 through the node N_M3, is arranged to mirror the current I_M3 to draw the current I_M2 from the node N_M1. Note that the current I_M3 can be related to the effect of process variations on the supply current I_S. For example, the current generator 302 or the oscillator 304 can be a MOS-based circuit. The process-dependent current I_PT can flow through a transistor of the current mirror 438 such that each of the process-dependent current I_PT and the current I_M3 can be related to a current response of the transistor to process variations.

[0056] In the example of FIG. 4, the current mirror 438 may include the transistors M₃₃ and M₃₄. Respective control terminals of the transistors M₃₃ and M₃₄ are coupled to a first connection terminal of the transistor M₃₃. Respective first connection terminals of the transistors M₃₃ and M₃₄ are coupled to the supply voltage VCC. Respective second connection terminals of the transistors M₃₃ and M₃₄ are coupled to the resistive element R3 and the node N_M3, respectively. Note that each of the transistors M₃₃ and M₃₄ can be implemented as a p-channel transistor. The process-dependent current I_PT can be related to the effect of process variations in the oscillator 304 (or the current generator 302) if the oscillator 304 (or the current generator 302) is PMOS (p-channel MOSFET) dominant.

[0057] A terminal of the resistive element R3 is coupled to the second connection terminal of the transistor M₃₃, and another terminal of the resistive element R3 is coupled to the reference voltage VSS. The process-dependent current I_PT can flow through a transistor of the current mirror 438 (e.g. the transistor M₃₃) and the resistive element R3. Note that the resistance of the resistive element R3 can change with temperature. Steering the process-dependent current I_PT to or from the resistive element R3 can achieve better temperature compensation. For example, the resistive element R3 can alleviate or prevent overcompensation of temperature variations.

[0058] As those skilled in the art can appreciate the operation of the oscillator circuit 400 shown in FIG. 4 after reading the above paragraphs directed to FIG. 1 to FIG. 3, further description is omitted here for brevity.

[0059] FIG. 5 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The oscillator circuit 500 may include the current generator 302 and the oscillator 304 shown in FIG. 3, and further include a current compensation circuit 510. The structure of the oscillator circuit 500 is similar/identical to that of the oscillator circuit 300 shown in FIG. 3 except for the current generator 536 included in the current generating circuit 530.

[0060] In the present embodiment, the control terminal and the first connection terminal of the transistor M₃₁ are coupled to the supply voltage VCC. The current mirror 338 can mirror the process-dependent current I_PT to provide the current I_M2. With the use of the current compensation circuit 510, the oscillator circuit 500 can achieve a high PSRR over different process corners. As those skilled in the art can appreciate the operation of the oscillator circuit 500 shown in FIG. 5 after reading the above paragraphs directed to FIG. 1 to FIG. 4, further description is omitted here for brevity.

[0061] FIG. 6 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The oscillator circuit 600 may include the current generator 302 and the oscillator 304 shown in FIG. 3, and further include a current compensation circuit 610. The structure of the oscillator circuit 600 is similar/identical to that of the oscillator circuit 400 shown in FIG. 4 except for the current generator 636 included in the current generating circuit 630.

[0062] In the present embodiment, the second connection terminal of the transistor M₃₃ are coupled to the reference voltage VSS. The current mirror 438 can mirror the process-dependent current I_PT to provide the current I_M3. With the use of the current compensation circuit 610, the oscillator circuit 600 can achieve a high PSRR over different process corners. As those skilled in the art can appreciate the operation of the oscillator circuit 600 shown in FIG. 6 after reading the above paragraphs directed to FIG. 1 to FIG. 5, further description is omitted here for brevity.

[0063] FIG. 7 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The structure of the oscillator circuit 700 is similar/identical to that of the oscillator circuit 300 shown in FIG. 3 except for the current compensation circuit 710. In the present embodiment, the input current I_IN injected to the node N_M1 can serve as the current I_M1 inputted to the transistor M₄₁. With the use of the current compensation circuit 710, the oscillator circuit 700 can achieve a high PSRR. As those skilled in the art can appreciate the operation of the oscillator circuit 700 shown in FIG. 7 after reading the above paragraphs directed to FIG. 1 to FIG. 6, further description is omitted here for brevity.

[0064] FIG. 8 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The structure of the oscillator circuit 800 is similar/identical to that of the oscillator circuit 300 shown in FIG. 3 except for the current compensation circuit 810. In the present embodiment, the input signal S_IN shown in FIG. 2 can be a voltage signal applied to the node N_M1. For example, the signal generator 820 can be configured to provide an input voltage V_IN by capacitively coupling the supply voltage VCC to the node N_M1. The input voltage V_IN can serve as an embodiment of the input signal S_IN shown in FIG. 2.

[0065] The signal generator 820 can be implemented using a voltage source arranged for providing the input voltage V_IN. In the example of FIG. 8, the signal generator 820 can include a capacitive element C1, which can act as a voltage source coupled between the supply node N_CG and the node N_M1. The input voltage V_IN can carry information on a variation in (or an alternating current (AC) component of) the supply voltage VCC.

[0066] The current generating circuit 830 may include, but is not limited to, a current source 834, a transistor M₈₁ and a transistor M₈₂. In the present embodiment, the current source 834 can be related to the current generator 302. For example, the current source 834 can be coupled to the supply voltage VCC through the node N_M2, and arranged to provide the process-dependent current I_PT that can be related to the effect of process variations on the supply current I_S.

[0067] In addition, a connection terminal of the transistor M₈₁ is coupled to a control terminal of the transistor M₈₁, and is further coupled to the current source 834 to receive the process-dependent current I_PT. A connection terminal of the transistor M₈₂ is coupled to the node N_C1. A control terminal of the transistor M₈₂ is coupled to the node N_M1, and electrically connected to the control terminal of the transistor M₈₁. In the example of FIG. 8, the control terminal of the transistor M₈₂ can be electrically connected to the control terminal of the transistor M₈₁ through a resistive element R4.

[0068] In operation, the current generating circuit 830 can act as a current mirror that mirrors the process-dependent current I_PT to provide a first portion of the compensation current I_C (represented by the current I_CP). A first component of the node voltage V_N (e.g. a direct current (DC) component) can be developed at the node N_M1 in response to the process-dependent current I_PT. Additionally, supply noise or voltage ripples appearing on the supply voltage VCC can be capacitively coupled to the node N_M1, thereby producing the input voltage V_IN which can serve as a second component of the node voltage V_N (e.g. an AC component). A second portion of the current I_C (represented by the current I_CV) can flow into the transistor M₈₂ in response to the second component of the node voltage V_N. Thus, the current I_C can be related to the effect of PVT variations. The current compensation circuit 810 can draw the current I_C from the supply node N_OSC to compensate for a variation in the supply current I_S. The oscillator 304 can generate the oscillating signal CK_OSC of a stable or substantially fixed frequency according to the driving current I_D.

[0069] With the use of the current compensation circuit 810, the oscillator circuit 800 can achieve a high PSRR over different process corners. Note that the circuit structure shown in FIG. 8 is provided for illustrative purposes, and is not intended to limit the scope of the present disclosure. In some embodiments, the first component of the node voltage V_N can be developed using other circuits capable of providing the process-dependent current I_PT. In some embodiments, the resistive element R4 may be optional.

[0070] FIG. 9 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The structure of the oscillator circuit 900 is similar/identical to that of the oscillator circuit 800 shown in FIG. 8 except for the current generating circuit 930. In the present embodiment, the current generating circuit 930 includes, but is not limited to, the current generator 336 shown in FIG. 3 and a current mirror 932. The current mirror 932 may include the resistive element R4 and the transistors M₈₁ and M₈₂ shown in FIG. 8, and a current source 934 arranged for providing a reference current I_REF. A current level of the reference current I_REF can be independent of or insensitive to a variation in the supply voltage VCC.

[0071] The current mirror 932 can be configured to mirror a first portion of the reference current I_REF (i.e. the current Iri) to provide a first portion of the compensation current I_C (represented by the current I_C1). The current I_P1 can be related to process and/or temperature variations. For example, the current generator 336, coupled to the current mirror 932 through the node N_C2, is configured to draw a second portion of the reference current I_REF (i.e. the current I_P2) from the node N_C2 according to the process-dependent current I_PT. Both of the process-dependent current I_PT and the current I_P2 can be related to process and/or temperature variations. Thus, the current I_P1, a result of subtraction of the current I_P2 from the reference current I_REF, can be related to process and/or temperature variations.

[0072] In operation, the current mirror 932 mirrors the current I_P1 to provide the first portion of the compensation current I_C (i.e. the current I_C1 related to process and/or temperature variations). Supply noise or voltage ripples appearing on the supply voltage VCC can be capacitively coupled to the node N_M1, resulting in a second portion of the current I_C (represented by the current I_C2) flowing into the transistor M₈₂. The current I_C2 is related to voltage variations. Thus, the current I_C flowing into the node N_C1 can be related to the effect of PVT variations. The current compensation circuit 910 can draw the current I_C from the supply node N_OSC to compensate for a variation in the supply current I_S. The oscillator 304 can generate the oscillating signal CK_OSC of a stable or substantially fixed frequency according to the driving current I_D.

[0073] With the use of the current compensation circuit 910, the oscillator circuit 800 can achieve a high PSRR over different process corners, and have a low temperature coefficient. As those skilled in the art can appreciate the operation of the oscillator circuit 900 shown in FIG. 9 after reading the above paragraphs directed to FIG. 1 to FIG. 8, further description is omitted here for brevity.

[0074] FIG. 10 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The structure of the oscillator circuit 1000 is similar/identical to that of the oscillator circuit 900 shown in FIG. 9 except that the current compensation circuit 1010 utilizes the current generator 436 shown in FIG. 4 to compensate for process and temperature variations. In some embodiments, the resistive element R3 may be optional. As those skilled in the art can appreciate the operation of the oscillator circuit 1000 shown in FIG. 10 after reading the above paragraphs directed to FIG. 1 to FIG. 9, further description is omitted here for brevity.

[0075] FIG. 11 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The structure of the oscillator circuit 1100 is similar/identical to that of the oscillator circuit 900 shown in FIG. 9 except for the current compensation circuit 1110. In the present embodiment, the current compensation circuit 1110 includes the signal generator 820 shown in FIG. 8, and a current generating circuit 1130.

[0076] The current generating circuit 1130 includes, but is not limited to, a current mirror 932 shown in FIG. 9 and a current generator 1136. In the example of FIG. 11, the current generator 1136 includes the current mirror 438 and the resistive element R3 shown in FIG. 4. The current mirror 438, coupled to the node N_C2, is arranged to mirror the process-dependent current I_PT to provide a current I_P3 flowing into the node N_C2. The current I_P3 can be related to the effect of process variations and/or temperature variations on the supply current I_S. The current mirror 932 can provide a first portion of the compensation current I_C (related to process and/or temperature variations) by mirroring the current I_P4, i.e. a sum of the reference current I_REF and the current I_P3. The current mirror 932 can further provide a second portion of the compensation current I_C in response to power supply noise capacitively coupled to the node N_M1. Thus, the current I_C flowing into the node N_C1 can be related to the effect of PVT variations.

[0077] With the use of the current compensation circuit 1110, the oscillator circuit 1000 can achieve a high PSRR over different process corners, and have a low temperature coefficient. As those skilled in the art can appreciate the operation of the oscillator circuit 1100 shown in FIG. 11 after reading the above paragraphs directed to FIG. 1 to FIG. 10, further description is omitted here for brevity.

[0078] FIG. 12 illustrates an implementation of the oscillator circuit 100 shown in FIG. 1 in accordance with some embodiments of the present disclosure. The structure of the oscillator circuit 1200 is similar/identical to that of the oscillator circuit 800 shown in FIG. 8 except for the current generating circuit 1230. In the present embodiment, the node N_M2 of the current generating circuit 1230 is coupled to the supply node N_OSC to determine the current I_C. The current generating circuit 1230 may include, but is not limited to, a transistor M₁₂₁ and a resistive element R5. A control terminal of the transistor M₁₂₁ is coupled to the node N_M1, a first connection terminal of the transistor M₁₂₁ is coupled to the supply node N_OSC through the node N_M2, and a second connection terminal of the transistor M₁₂₁ is coupled to the reference voltage VSS. The resistive element R5 is arranged for coupling the first connection terminal of the transistor M₁₂₁ to the control terminal of the transistor M₁₂₁.

[0079] In operation, when the supply voltage VCC stays at a constant level, the resistive element R5 coupled between the first connection terminal and the control terminal of the transistor M₁₂₁ can enable the process-dependent current I_PT to flow into the transistor M₁₂₁ through the node N_M2. The process-dependent current I_PT can be related to the effect of process variations on the supply current I_S. In addition, when supply noise or voltage ripples appear on the supply voltage VCC, the signal generator 820 can capacitively couple the supply noise or voltage ripples to the node N_M1 to thereby produce the input voltage V_IN, and the current compensation circuit 1210 can draw the current I_C from the supply node N_OSC to compensate for a variation in the supply current I_S. The oscillator 304 can generate the oscillating signal CK_OSC of a stable or substantially fixed frequency according to the driving current I_D. In the example of FIG. 12, the current I_C includes the process-dependent current I_PT and a current I_P0, which flows into the transistor M₁₂₁ through the node N_M2 in response to the input voltage V_IN.

[0080] Note that at least a portion of the compensation current provided by the proposed current compensation scheme can be generated according to a supply voltage and a reference signal. At least the portion of the compensation current may correspond to the effect of supply noise or voltage variations on the supply current. A signal level of the reference signal is independent of a variation in the supply voltage.

[0081] Referring again to FIG. 2, the current compensation circuit 110 can be configured to generate at least a portion of the current I_C according to the supply voltage VCC and a reference signal S_REF. A signal level of the reference signal S_REF is independent of a variation in the supply voltage VCC, and at least the portion of the current I_C changes in response to a variation in the supply voltage VCC. By way of example but not limitation, the reference signal S_REF can be a reference voltage that is unaffected by the variation in the supply voltage VCC. As another example, the reference signal S_REF can be a reference current that is unaffected by the variation in the supply voltage VCC. In addition, the current compensation circuit 110 can be configured to generate another portion of the current I_C according to a process-dependent current (e.g. the process-dependent current I_PT).

[0082] For example, referring again to FIG. 3, the reference signal S_REF shown in FIG. 2 can be implemented using the node voltage V_N coupled to the node N_C2. The signal generator 320 and the current mirror 332 can serve as at least a portion of a current generator 316. The current generator 316 can be configured to generate the input current I_IN according to the supply voltage VCC and the node voltage V_N, and draw the current I_C from the supply node N_OSC according to the input current I_IN and a process-dependent current applied to the node N_M1 (i.e. the current I_M2). In addition, the current generator 336 is configured to generate the process-dependent current I_PT, and provide the current I_M2 according to the process-dependent current I_PT. The combination of the input current I_IN and the current I_M2, e.g. the subtraction of the current I_M2 from the input current I_IN, can serve as a compensation current generated by the compensation circuit 310. The compensation circuit 310 can draw a compensation current (i.e. the current I_C) from the supply node N_OSC according to the combination of the input current I_IN and the current I_M2.

[0083] As another example, referring again to FIG. 8, the reference signal S_REF shown in FIG. 2 can be implemented using the reference voltage VSS, which has a voltage level independent of a variation in the supply voltage VCC. The capacitive element C1 and the transistor M₈₂ can serve as at least a portion of a current generator 816. The current generator 816 can be configured to generate an input current (i.e. the current I_CV) according to the supply voltage VCC and the reference voltage VSS, and draw the current I_C from the supply node N_OSC according to the input current and a first component (e.g. the DC component) of the node voltage V_N at the node N_M1. Additionally or alternatively, the resistive element R4, the current source 834 and the transistor M₈₁ can serve as at least a portion of a current generator 836. The current generator 836 can be configured to generate the process-dependent current I_PT, and provide the first component (e.g. the DC component) of the node voltage V_N according to the process-dependent current I_PT. The combination of the current I_CP (i.e. a portion of the current I_C that flows into the transistor M₈₂ in response to the first component (e.g. the DC component) of the node voltage V_N) and the current I_CV (i.e. a portion of the current I_C that flows into the transistor M₈₂ in response to the second component (e.g. the AC component) of the node voltage V_N) can serve as the compensation current provided by the compensation circuit 810 (i.e. the current I_C).

[0084] As still another example, referring again to FIG. 9, the reference signal S_REF shown in FIG. 2 can be implemented using the reference voltage VSS. The signal generator 820 and the current mirror 932 can serve as at least a portion of a current generator 916. The current generator 916 can be configured to generate the current I_C1 according to a subtraction of the current I_P2 (i.e. a process-dependent current) from the reference current I_REF. The current generator 916 can be further configured to generate the current I_C2 according to the supply voltage VCC and the node voltage V_N.

[0085] The circuit structures described above are provided for illustrative purposes, and are not intended to limit the scope of the present disclosure. For example, in some embodiments, the current compensation circuit 110 shown in FIG. 2 can be implemented to include a first current generator and a second current generator. The first current generator can be configured to generate an input current (e.g. the input current I_IN shown in FIG. 3, or the current I_CV shown in FIG. 8) according to the supply voltage VCC and the reference signal S_REF (e.g. the node voltage V_N shown in FIG. 3, or the reference voltage VSS in FIG. 8), and draw the current I_C from the supply node N_OSC according to the input current and an electrical signal (e.g. the current I_M2 shown in FIG. 3, or the DC component of the node voltage V_N shown in FIG. 8) applied to the node N_M1. The first current generator and/or the second current generator can be implemented using other circuit structures without departing from the scope of the present disclosure.

[0086] In addition, referring again to FIG. 12, the reference signal S_REF shown in FIG. 2 can be implemented using the reference voltage VSS that has a voltage level independent of a variation in the supply voltage VCC. The capacitive element C1 is arranged to couple the supply voltage VCC to the control terminal of the transistor M₁₂₁ to enable a first portion of the current I_C (i.e. the current I_P0) to flow into the transistor M₁₂₁. The resistive element R5 is arranged to couple the first connection terminal of the transistor M₁₂₁ to the control terminal of the transistor M₁₂₁ to enable a second portion of the current I_C (i.e. the process-dependent current I_PT) to flow into the transistor M₁₂₁.

[0087] As those skilled in the art can appreciate the operation of a current compensation circuit, which is configured to generate a first portion of a compensation current according to a supply voltage and a reference signal and generate a second portion of the compensation current according to a process-dependent current, after reading the above paragraphs directed to FIG. 1 to FIG. 12, further description is omitted here for brevity.

[0088] With the use of the proposed current compensation scheme, an oscillator circuit can generate an oscillating signal of a stable or substantially fixed frequency under power supply variations. The proposed current compensation scheme can enable the oscillator circuit to enhance the immunity to power supply noise. In addition, the oscillator circuit can be utilized to implement a PLL with reduced jitter and phase noise. The proposed current compensation scheme can improve the PSRR performance without increasing the voltage headroom of the PLL.

[0089] As used herein, the terms “substantially” are used to describe and account for small variations. When used in conduction with an event or circumstance, the terms can refer to instances in which the event of circumstance occurs precisely as well as instances in which the event or circumstance occurs to a close approximation. As used herein with respect to ta given value or range, the term “substantially” generally means within ±10%, ±5%, ±1%, or ±0.5% of the given value or range. Ranges can be expressed herein as from one endpoint to another endpoint or between two endpoints. All ranges disclosed herein are inclusive of the endpoints, unless specified otherwise. In addition, when referring to numerical values or characteristics as “substantially” the same, the term can refer to the values lying within ±10%, ±5%, ±1%, or ±0.5% of an average of the values.

[0090] The foregoing outlines features of several embodiments so that those skilled in the art may better understand the aspects of the present disclosure. Those skilled in the art should appreciate that they may readily use the present disclosure as a basis for designing or modifying other processes and structures for carrying out the same purposes and/or achieving the same advantages of the embodiments introduced herein. Those skilled in the art should also realize that such equivalent constructions do not depart from the spirit and scope of the present disclosure, and that they may make various changes, substitutions, and alterations herein without departing from the spirit and scope of the present disclosure.